Power supply device, driving device, control method, and storage medium

ABSTRACT

A power supply device is provided with an output unit, an input unit, a rectifier circuit, a switching circuit, and a controller. The input unit inputs an AC input. The rectifier circuit has a smoothing capacitor and converts the AC input into a rectified output. The switching circuit switches between an ON-state in which input impedance is low and an OFF-state in which input impedance is higher than that in the ON-state. The controller sets a modulation ratio such that, when controlling the switching circuit to switch between the ON-state and the OFF-state, at least a portion of a period, in which the modulation width of the modulation duty ratio becomes maximum, is included in a period from a time when an input current inputted to the smoothing capacitor is generated to a time when the voltage of the smoothing capacitor reaches a maximum value.

TECHNICAL FIELD

The present invention relates to a power supply device converting an ACinput into a predetermined output and outputting the predeterminedoutput to a load, a driving device using a motor as the load of thepower supply device, a control method for the power supply device, and aprogram causing a computer to perform the control method.

BACKGROUND ART

In the related art, power supply devices that convert AC inputs into DCoutputs and output the DC outputs are known. The power supply devicesinclude a switching element that switches between an ON-state and anOFF-state at a predetermined period and can adjust the power of an DCoutput to be supplied to a load driven by the power supply device byadjusting a ratio (referred to as a duty ratio) of a period in which theswitching element is in the ON-state (or the OFF-state) with respect tothe predetermined period. A power supply device converting an AC inputinto a DC output includes a capacitor maintaining a DC output at aconstant voltage (referred to as a smoothing capacitor) (for example,see Patent Literature 1).

CITATION LIST Patent Literature [Patent Literature 1]

Japanese Patent Laid-Open No. H11-179559

SUMMARY OF INVENTION Technical Problem

In the power supply device, for example, when a current is not suppliedfrom a rectifier circuit to the smoothing capacitor and the smoothingcapacitor is not charged, the smoothing capacitor is discharged, andthus a voltage of a DC output is lowered. On the other hand, when acurrent is supplied from the rectifier circuit to the smoothingcapacitor, the smoothing capacitor is charged. Thus, the voltage of theDC output lowered due to the discharging can be raised again.

That is, in the power supply device, the voltage (an average of thevoltages) of the DC output can be constant when discharging and chargingof the smoothing capacitor is periodically repeated.

On the other hand, the repeated discharging and charging of thesmoothing capacitor during running of the power supply device means thata current always flows in the smoothing capacitor during running of thepower supply device. As a result, the smoothing capacitor deteriorateswhenever the power supply device is used and the deterioration of thesmoothing capacitor causes shortening of a lifespan of the power supplydevice.

An objective of the present invention is to suppress deterioration of asmoothing capacitor provided in a power supply device.

Solution to Problem

According to an exemplary embodiment of the present specification, apower supply device includes an output unit, an input unit, a rectifiercircuit, a switching circuit, and a controller. The output unit connectsa load. The input unit inputs an AC input varying at a predeterminedperiod between a positive voltage and a negative voltage. The rectifiercircuit is a circuit that converts the AC input from the input unit intoa rectified output which is one of the positive voltage and the negativevoltage and includes a smoothing capacitor that smoothes the rectifiedoutput. The switching circuit connects the smoothing capacitor as aninput, connects the output unit as an output, and switches between anON-state in which input impedance viewed from the smoothing capacitor islow and an OFF-state in which the input impedance is higher than theinput impedance in the ON-state at a switching period shorter than thepredetermined period of the AC input.

The controller sets a modulation duty ratio such that at least a portionof a period in which a modulation width of the modulation duty ratiobecomes a maximum value is included in a period from a time at which aninput current input to the smoothing capacitor is generated to a time atwhich a voltage of the smoothing capacitor becomes a maximum when theswitching circuit is controlled to switch between the ON-state and theOFF-state while outputting a modulation duty ratio at which a duty ratiowhich is a ratio of a period in which the ON-state is maintained to theswitching period is modulated in accordance with a variation in therectified output.

Advantageous Effects of Invention

In the power supply device according to an exemplary embodiment of thepresent specification, by further lengthening a state in which inputimpedance viewed from the smoothing capacitor is low within a periodfrom a time at which an input current input to the smoothing capacitoris generated to a time at which the voltage in the smoothing capacitorbecomes a maximum, that is, a period in which the smoothing capacitor ischarged, it is possible to reduce a current amount flowing in thesmoothing capacitor and suppress deterioration of the smoothingcapacitor.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a diagram illustrating a configuration of a driving deviceaccording to a first embodiment.

FIG. 2 is a diagram illustrating an example of a voltage waveform of anAC input.

FIG. 3A is a diagram schematically illustrating full-wave rectification.

FIG. 3B is a diagram schematically illustrating half-wave rectification.

FIG. 4 is a diagram illustrating an example of a voltage waveform of arectified output.

FIG. 5 is a diagram illustrating a detailed configuration of the drivingdevice.

FIG. 6 is a diagram illustrating an example of a time chart illustratingan operation of each switching element of a switching circuit.

FIG. 7 is a flowchart illustrating a control method for the drivingdevice (power source device) according to a first embodiment.

FIG. 8 is a diagram illustrating an example of a variation in amodulation duty ratio set in the driving device.

FIG. 9 is a diagram illustrating an example of a load current inaccordance with presence or absence of delay of the modulation dutyratio.

FIG. 10 is a diagram illustrating an example of dependency of a delaytime of the modulation duty ratio of a ripple voltage and a ripplecurrent.

FIG. 11 is a diagram illustrating an example of a variation in themodulation duty ratio and a load current when an excessive delay time isset.

FIG. 12 is a diagram illustrating an example of a basic duty ratio setin each switching element.

FIG. 13 is a diagram illustrating an example of a modulation duty ratioof each switching element.

FIG. 14 is a diagram illustrating an example of a modulation duty ratiowhen a variation frequency of the basic duty ratio is more increased.

FIG. 15 is a diagram illustrating an example of a modulation duty ratiowhen a variation frequency of the basic duty ratio is further moreincreased.

FIG. 16 is a diagram illustrating another example of the basic dutyratio set in each switching element.

FIG. 17 is a diagram illustrating other examples of a modulation widthof a modulation duty ratio and the modulation duty ratio of eachswitching element.

FIG. 18 is a diagram illustrating a modulation width of the modulationduty ratio and another example of the modulation duty ratio when thevariation frequency of the basic duty ratio is more increased.

FIG. 19 is a diagram illustrating a modulation width of the modulationduty ratio and another example of the modulation duty ratio when thevariation frequency of the basic duty ratio is further more increased.

FIG. 20 is a diagram illustrating an example of a time chartschematically illustrating advance angle modulation.

FIG. 21 is a flowchart illustrating a control method for a drivingdevice according to a third embodiment.

FIG. 22 is a flowchart illustrating a method of calculating an advanceangle.

FIG. 23 is a diagram illustrating an example of a variation in theadvance angle set in the driving device.

FIG. 24 is a diagram illustrating an example of a load current inaccordance with presence or absence of advance angle modulation.

FIG. 25 is a diagram illustrating an example of a load current inaccordance with presence or absence of delay of advance anglemodulation.

FIG. 26 is a diagram illustrating an example of dependency of a delaytime of an advance angle of a ripple current.

FIG. 27 is a diagram illustrating an example of dependency of a delaytime of an advance angle at a maximum value of a load current.

FIG. 28 is a diagram illustrating another embodiment (part 1) of amethod of calculating a duty ratio/advance angle.

FIG. 29 is a diagram illustrating another embodiment (part 2) of themethod of calculating the duty ratio/advance angle.

FIG. 30 is a diagram illustrating another embodiment (part 3) of themethod of calculating the duty ratio/advance angle.

DESCRIPTION OF EMBODIMENTS

Hereinafter, embodiments of the present invention will be described withreference to the drawings. The scope of the present invention is notlimited to the following embodiments and can be arbitrarily changedwithin the scope of the technical spirit of the present invention.

In the following description “a switching element being ON” means thatboth ends of a switching element are electrically connected to eachother or a switching element is in a low impedance state. On the otherhand, “a switching element being OFF” means that both ends of aswitching element are electrically disconnected from each other or aswitching element is in a high impedance state.

First Embodiment 1-1. Configuration of Driving Device

FIG. 1 is a diagram illustrating a configuration of a driving device 100according to a first embodiment. The driving device 100 includes a powersupply device 1 and a load LO. The power supply device 1 converts ACpower into a predetermined electrical output (a high-frequency output)and outputs the electrical output to the load LO. The load LO is, forexample, a motor, an induction heating (IH) device, or the like. Themotor is, for example, a brushless motor such as a synchronous motor oran induction motor.

The driving device 100 using the load LO as a motor can be used as, forexample, a suction device of a cleaner, a driving device of an electrictool, a driving device of a vehicle, or the like. The driving device 100using the load LO as an induction heating device can be, for example, adriving device or like of an IH cooker.

1-2. Overall Configuration of Power Supply Device

As illustrated in FIG. 1, the power supply device 1 includes an inputunit 11, an output unit 13, a rectifier circuit 15, a switching circuit17, and a controller 19.

The input unit 11 is a connection terminal connecting an AC power supplyPS to the rectifier circuit 15. In the embodiment, the AC power supplyPS is a single-phase AC power supply that has two poles. The AC powersupply PS outputs an AC input V_(in) varying in a sinusoidal waveform ata predetermined frequency between positive and negative voltages, asillustrated in FIG. 2. When the AC power supply PS is a commercial powersupply, the predetermined frequency is, for example, 50 Hz or 60 Hz.

The AC power supply PS connected to the input unit 11 is, for example, agenerally supplied household or commercial AC power supply, inverterpower supply, an AC generator, or the like. The input unit 11 mayconnect the AC power supply PS via a transformer (not illustrated). Inthis case, the input unit 11 inputs an AC input V_(in) lower or higherthan a voltage output from the AC power supply PS.

The output unit 13 is a connection terminal connecting the load LOdriven in the driving device 100 to the switching circuit 17.

The rectifier circuit 15 includes a rectifier unit 151 and a smoothingcapacitor SC. The rectifier unit 151 is a circuit that converts the ACinput V_(in) input from the input unit 11 into a rectified outputV_(out) which is one of positive and negative voltages. As will bedescribed below, the rectifier unit 151 mainly includes rectifierelements.

As will be described below, the rectifier unit 151 according to theembodiment is a full-wave rectifier circuit that passes a positive-sidevoltage of the AC input V_(in) and inverts a negative-side voltage intoa positive-side voltage and passes the inverted positive-side voltage,as illustrated in FIG. 3A. Thus, the rectifier unit 151 outputs arectified output V_(out) that has a positive voltage across a period ofan AC period (a predetermined period) of the AC input V_(in).

In another embodiment, conversely, the rectifier unit 151 may be afull-wave rectifier circuit that passes a negative-side voltage of theAC input V_(in) and inverts a positive-side voltage into a negative-sidevoltage and passes the inverted negative-side voltage. In this case, therectifier unit 151 outputs a rectified output V_(out) that has anegative voltage across a period of an AC period (a predeterminedperiod) of the AC input V_(in).

In still another embodiment, the rectifier unit 151 may be a half-waverectifier circuit that passes one of a positive-side voltage and anegative-side voltage of the AC input V_(in) and does not pass the othervoltage, as illustrated in FIG. 3B. Thus, the rectifier unit 151 outputsa rectified output V_(out) that has positive or negative voltage duringonly half of a period of the AC period of the AC input V_(in).

The smoothing capacitor SC is a capacitor that “smoothes” the rectifiedoutput V_(out). Specifically, for example, the smoothing capacitor SCsets a voltage output obtained by rectifying the AC input V_(in), asillustrated in FIG. 3A, to the positive side of the voltage as therectified output V_(out) that has an average voltage V_(ave) andperiodically varies near the average voltage V_(ave), as illustrated inFIG. 4. A voltage periodically varying near the average voltage V_(ave)is also called a “ripple voltage”. A voltage of the smoothing capacitorSC corresponds to the rectified output V_(out).

The “ripple voltage” is mainly produced by repeating charging anddischarging of the smooth capacitor SC and a variation width isdetermined in accordance with a time constant that the smoothingcapacitor SC particularly has. Accordingly, a capacitor that has largecapacitance (a large time constant) such as an electrolytic capacitor isused as, for example, the smoothing capacitor SC. Thus, a variationwidth of the “ripple voltage” decreases and the rectified output V_(out)that has an almost constant voltage at the average voltage V_(ave) canbe output.

An input of the switching circuit 17 is connected in parallel to thesmoothing capacitor SC. The switching circuit 17 is mainly a circuitthat includes a switching element of which ON and OFF is controlled bythe controller 19. By controlling ON and OFF of the switching element,the switching circuit 17 switches between an ON-state in which inputimpedance is low when viewed from the smoothing capacitor SC and anOFF-state in which the input impedance is higher than in the ON-state.

The input impedance is defined as impedance from the smoothing capacitorSC that includes the switching circuit 17, the output unit 13 connectingan output of the switching circuit 17, and the load LO connected to theoutput unit 13.

The switching circuit 17 switches between an ON-state and an OFF-stateat a switching period shorter than an AC period of the AC input V_(in).As will be described below, the switching circuit 17 according to theembodiment is an inverter that outputs an AC output that has power withany frequency and any magnitude to the load LO by appropriatelyadjusting a switching pattern of the ON-state and the OFF-state.

In another embodiment, the switching circuit 17 may be, for example, aboosting chopper type converter, a step-down chopper type converter, anLLC converter, a pseudo-resonant flyback converter, or the like. In theswitching circuit 17, power to be input to the switching circuit 17 canbe adjusted in accordance with a frequency of pulse width modulation(PWM) control of turning ON and OFF of the switching element included insuch a circuit.

The controller 19 is a system that controls the power supply device 1.The controller 19 particularly controls switching between the ON-stateand the OFF-state of the switching circuit 17. Accordingly, thecontroller 19 can be configured as hardware including a PWM signalgeneration circuit, a potential measurement circuit, and/or a currentmeasurement circuit.

Alternatively, the controller 19 may be a computer system that includesa CPU (central processing unit), a storage element such as a RAM (randomaccess memory) or a ROM (read only memory), an A/D interface(analog/digital interface), and a D/A interface (digital/analoginterface). In this case, a program that is stored in the storageelement or the like and can be executed by the controller 19 may realizecontrol of the power supply device 1 performed by the controller 19.

Additionally, a system on chip (SoC), an application specific integratedcircuit (ASIC), or the like in which the PWM signal generation circuit,the potential measurement circuit, and/or the current measurementcircuit are formed on one chip can be used as the controller 19.

1-3. Detailed Configuration of Driving Device

Hereinafter, a detailed configuration of the driving device 100according to the first embodiment will be described with reference toFIG. 5. The driving device 100 according to the first embodiment is thedriving device 100 in which a three-phase brushless motor M is used asthe load LO, as illustrated in FIG. 5. The three-phase brushless motor Mhas three phases which are U, V, and W phases.

Accordingly, in the driving device 100 according to the firstembodiment, three terminals connected to the respective phases of thethree-phase brushless motor M form the output unit 13. Hereinafter, adetailed configuration of the switching circuit 17 and the rectifierunit 151 of the rectifier circuit 15 will be described.

1-3-1. Detailed Configuration of Rectifier Unit

In the first embodiment described above, the rectifier unit 151 is afull-wave rectifier circuit. Specifically, as illustrated in FIG. 5, therectifier unit 151 includes four rectifier elements D1 to D4 and aninductor element L.

The four rectifier elements D1 to D4 form a bridge circuit.Specifically, the rectifier element D1 connects an anode side to a firstinput terminal I1 of the input unit 11 via the inductor element L andconnects a cathode side to one end of the smoothing capacitor SC. Therectifier element D2 connects the cathode side to the first inputterminal I1 via the inductor element L and connects the anode side tothe other end of the smoothing capacitor SC. The other end of thesmoothing capacitor SC is an opposite side to the one end to which thecathode side of the rectifier element D1 is connected.

The rectifier element D3 connects the anode side to a second inputterminal I2 and connects the cathode side to one end of the smoothingcapacitor SC. The rectifier element D4 connects the cathode side to thesecond input terminal I2 and connects the anode side to the other end ofthe smoothing capacitor SC.

In the bridge circuit formed by the four rectifier elements D1 to D4,the rectifier elements D1 and D4 are in a conductive state and therectifier element D2 and D3 are in a nonconductive state when thevoltage of the AC input V_(in) is positive and the absolute value of thevoltage exceeds a voltage of the smoothing capacitor SC. In theembodiment, the voltage of the AC input V_(in) is assumed to be positivewhen a potential of the first input terminal I1 is higher than apotential of the second input terminal I2.

As a result, a voltage of the smoothing capacitor SC is a positivevoltage. That is, the one end of the smoothing capacitor SC has apositive potential and the other side has a negative potential (0potential).

On the other hand, the rectifier elements D2 and D3 are in a conductivestate and the rectifier elements D1 and D4 are in a nonconductive statewhen the voltage of the AC input V_(in) is negative and the absolutevalue of the voltage exceeds the voltage of the smoothing capacitor SC.As a result, one end of the smoothing capacitor SC has a positivepotential and the other end has a negative potential (0 potential). Thatis, the voltage of the smoothing capacitor SC is a positive voltage.

For example, diodes such as PN diodes or Schottky diodes can be used asthe rectifier elements D1 to D4.

The inductor element L is, for example, an example, an inductancecomponent such as a coil. The inductor element L connects one end to thefirst input terminal I1 and connects the other end to the anode side ofthe rectifier element D1 or the cathode side of the rectifier elementD2. The inductor element L connected in this way forms a passive powerfactor improvement circuit along with the smoothing capacitor SC andsuppresses occurrence of harmonics in an input current from the AC powersupply PS.

In the foregoing example, the inductor element L is provided between thefirst input terminal I1 and the rectifier elements D1 and D2, but thepresent invention is not limited thereto. For example, the inductorelement L may be provided between the second input terminal I2 and therectifier elements D3 and D4 or may be provided in both thereof. Therectifier elements may be disposed on the side of the input unit 11 withrespect to the inductor element L.

1-3-2. Detailed Configuration of Switching Circuit

Since the load LO of the driving device 100 is the three-phase brushlessmotor M, the switching circuit 17 (an example of a motor controlcircuit) is an inverter circuit. Specifically, the switching circuit 17includes six switching elements SW1 to SW6 and six rectifier elements D5to D10 corresponding to the switching elements SW1 to SW6.

The switching element SW1 connects one end to one end of the smoothingcapacitor SC and connects the other end to a U terminal of the outputunit 13. A control pole (gate pole) of the switching element SW1 isconnected to the controller 19. Thus, the switching element SW1 connectsor disconnects the one end of the smoothing capacitor SC to or from theU terminal under the control of the controller 19.

The rectifier element D5 corresponding to the switching element SW1connects one end of the anode side to the U terminal and connects theother end of the cathode side to one end of the smoothing capacitor SC.

The switching element SW2 connects one end to one end of the smoothingcapacitor SC and connects the other end to a V terminal of the outputunit 13. A control pole of the switching element SW2 is connected to thecontroller 19. Thus, the switching element SW2 connects or disconnectsthe one end of the smoothing capacitor SC to or from the V terminalunder the control of the controller 19.

The rectifier element D6 corresponding to the switching element SW2connects one end of the anode side to the V terminal and connects theother end of the cathode side to one end of the smoothing capacitor SC.

The switching element SW3 connects one end to one end of the smoothingcapacitor SC and connects the other end to a W terminal of the outputunit 13. A control pole of the switching element SW3 is connected to thecontroller 19. Thus, the switching element SW3 connects or disconnectsthe one end of the smoothing capacitor SC to or from the W terminalunder the control of the controller 19.

The rectifier element D7 corresponding to the switching element SW3connects one end of the anode side to the W terminal and connects theother end of the cathode side to one end of the smoothing capacitor SC.

The switching element SW4 connects one end to the U terminal of theoutput unit 13 and connects the other end to the other end of thesmoothing capacitor SC. A control pole of the switching element SW4 isconnected to the controller 19. Thus, the switching element SW4 connectsor disconnects the other end of the smoothing capacitor SC to or fromthe U terminal under the control of the controller 19.

The rectifier element D8 corresponding to the switching element SW4connects one end of the anode side to the other end of the smoothingcapacitor SC and connects the other end of the cathode side to the Uterminal.

The switching element SW5 connects one end to the V terminal of theoutput unit 13 and connects the other end to the other end of thesmoothing capacitor SC. A control pole of the switching element SW5 isconnected to the controller 19. Thus, the switching element SW5 connectsor disconnects the other end of the smoothing capacitor SC to or fromthe V terminal under the control of the controller 19.

The rectifier element D9 corresponding to the switching element SW5connects one end of the anode side to the other end of the smoothingcapacitor SC and connects the other end of the cathode side to the Vterminal.

The switching element SW6 connects one end to the W terminal of theoutput unit 13 and connects the other end to the other end of thesmoothing capacitor SC. A control pole of the switching element SW6 isconnected to the controller 19. Thus, the switching element SW6 connectsor disconnects the other end of the smoothing capacitor SC to or fromthe W terminal under the control of the controller 19.

The rectifier element D10 corresponding to the switching element SW6connects one end of the anode side to the other end of the smoothingcapacitor SC and connects the other end of the cathode side to the Wterminal.

Since the ON-state and the OFF-state are switched at a high speed in theswitching circuit 17, the switching elements SW1 to SW6 included in theswitching circuit 17 are preferably elements capable of performing aswitching operation at a high speed.

Accordingly, in the embodiment, the switching elements SW1 to SW6 are,for example, metal oxide semiconductor field effect transistors(MOSFETs). Additionally, for example, semiconductor elements that haveswitching characteristics, such as field effect transistors, bipolartransistors, insulated gate bipolar transistors (IGBTs), or thyristorsother than MOSFETs can be used. For the switching elements SW1 to SW6, aconfiguration of a combination of these semiconductor elements may beused.

The rectifier elements D5 to D10 are, for example, diodes such as PNdiodes or Schottky diodes. Additionally, the rectifier elements D5 toD10 may be rectifier elements formed in the corresponding switchingelements SW1 to SW6. The rectifier elements formed in the switchingelements SW1 to SW6 are called body diodes, parasitic diodes, or thelike.

Thus, in the switching circuit 17, the rectifier elements D5 to D10 maynot be separate components from the switching elements SW1 to SW6. As aresult, the number of components of the switching circuit 17 can bereduced.

In the switching circuit 17 that has the foregoing configuration, thecontroller 19 switches each of the switching elements SW1 to SW6 betweenON and OFF by applying an ON signal and an OFF signal of each of theswitching elements SW1 to SW6 to the control pole of each of theswitching elements SW1 to SW6. The ON signal is, for example, a positivevoltage signal. On the other hand, the OFF signal is, for example, azero voltage signal.

In the embodiment, a switching timing of each of the switching elementsSW1 to SW6 is determined based on 120-degree conduction scheme (to bedescribed below). Specifically, for example, as illustrated in FIG. 6,PWM control is performed on the switching elements SW1 to SW3 which arehigh-side switches.

Hereinafter, the three-phase brushless motor M serving as the load LOincludes, for example, a U-phase rotor detection element disposedbetween a W-phase winding and a U-phase winding, a V-phase rotordetection element disposed between the U-phase winding and a V-phasewinding, and a W-phase rotor detection element disposed between theV-phase winding and the W-phase winding (none of which is illustrated).The controller 19 can detect where the rotor of the three-phasebrushless motor M is from output signals of the rotor detectionelements.

As the rotor detection elements, for example, Hall elements detectingmagnetic fields which the rotors have can be used.

In the example illustrated in FIG. 6, a timing at which the U-phaserotor detection element detects passage of the rotor (in FIG. 6, a“U-phase” is turned on) is time 0 and a rotation period of the rotor isT. A voltage of the rectified output V_(out) is assumed to be positive.That is, a potential of a side connected to the switching element SW1 ofthe smoothing capacitor SC is assumed to be higher than a potential of aside connected to the switching element SW4.

During a time from 0 to T/6, the controller 19 turns on the switchingelements SW1 and SW5 and turns off the other switching elements. At thistime, the impedance of the switching element SW1, the U-phase, theV-phase, and the switching element SW5, from one end of the smoothingcapacitor SC to the other end of the smoothing capacitor SC is lowered.

As a result, the input impedance viewed from the smoothing capacitor SCis lowered and the rectified output V_(out) is supplied between theU-phase and the V-phase of the three-phase brushless motor M from therectifier circuit 15.

On the other hand, the impedance between one end of the smoothingcapacitor SC, and the U-phase, the V-phase and the W-phase is raised byturning off the switching element SW1 within that time range. As aresult, the input impedance viewed from the smoothing capacitor SC israised and the supply of the rectified output V_(out) between theU-phase and the V-phase from the rectifier circuit 15 is stopped. On theother hand, a reflux current flows along a path of the rectifier elementD8, the U-phase, the V-phase, and the switching element SW5.

During a time from T/6 to 2T/6 (=T/3), the controller 19 turns on theswitching elements SW1 and SW6 and turns off the other switchingelements. At this time, the impedance of the switching element SW1, theU-phase, the W-phase, and the switching element SW6, from one end of thesmoothing capacitor SC to the other end of the smoothing capacitor SC islowered.

Thus, the input impedance viewed from the smoothing capacitor SC islowered and the rectified output V_(out) is supplied between the U-phaseand the W-phase of the three-phase brushless motor M from the rectifiercircuit 15.

On the other hand, the impedance between one end of the smoothingcapacitor SC, and the U-phase, the V-phase and the W-phase is raised byturning off the switching element SW1 within that time range. As aresult, the input impedance viewed from the smoothing capacitor SC israised and the supply of the rectified output V_(out) between theU-phase and the W-phase from the rectifier circuit 15 is stopped. On theother hand, a reflux current flows along a path of the rectifier elementD8, the U-phase, the W-phase, and the switching element SW6.

From T/3 to 3T/6 (=T/2) which is a timing at which the V-phase rotordetection element is turned on, the controller 19 turns on the switchingelements SW2 and SW6 and turns off the other switching elements. At thistime, the impedance of the switching element SW2, the V-phase, theW-phase, and the switching element SW6, from one end of the smoothingcapacitor SC to the other end of the smoothing capacitor SC is lowered.

As a result, the input impedance viewed from the smoothing capacitor SCis lowered and the rectified output V_(out) is supplied between theV-phase and the W-phase of the three-phase brushless motor M from therectifier circuit 15.

On the other hand, the impedance between one end of the smoothingcapacitor SC, and the U-phase, the V-phase, and the W-phase is raised byturning off the switching element SW2 within that time range. As aresult, the input impedance viewed from the smoothing capacitor SC israised and the supply of the rectified output V_(out) between theV-phase and the W-phase from the rectifier circuit 15 is stopped. On theother hand, a reflux current flows along a path of the rectifier elementD9, the V-phase, the W-phase, and the switching element SW6.

During a time from T/2 to 4T/6 (=2T/3), the controller 19 turns on theswitching elements SW2 and SW4 and turns off the other switchingelements. At this time, the impedance of the switching element SW2, theV-phase, the U-phase, and the switching element SW4, from one end of thesmoothing capacitor SC to the other end of the smoothing capacitor SC islowered.

As a result, the input impedance viewed from the smoothing capacitor SCis lowered and the rectified output V_(out) is supplied between theV-phase and the U-phase of the three-phase brushless motor M from therectifier circuit 15.

On the other hand, the impedance between one end of the smoothingcapacitor SC, and the U-phase, the V-phase, and the W-phase is raised byturning off the switching element SW2 within that time range. As aresult, the input impedance viewed from the smoothing capacitor SC israised and the supply of the rectified output V_(out) between theV-phase and the U-phase from the rectifier circuit 15 is stopped. On theother hand, a reflux current flows along a path of the rectifier elementD9, the V-phase, the U-phase, and the switching element SW4.

From 2T/3 to 5T/6 which is a timing at which the W-phase rotor detectionelement is turned on, the controller 19 turns on the switching elementsSW3 and SW4 and turns off the other switching elements. At this time,the impedance of the switching element SW3, the W-phase, the U-phase,and the switching element SW4, from one end of the smoothing capacitorSC to the other end of the smoothing capacitor SC is lowered.

As a result, the input impedance viewed from the smoothing capacitor SCis lowered and the rectified output V_(out) is supplied between theW-phase and the U-phase of the three-phase brushless motor M from therectifier circuit 15.

On the other hand, the impedance between one end of the smoothingcapacitor SC, and the U-phase, the V-phase, and the W-phase is raised byturning off the switching element SW3 within that time range. As aresult, the input impedance viewed from the smoothing capacitor SC israised and the supply of the rectified output V_(out) between theW-phase and the U-phase from the rectifier circuit 15 is stopped. On theother hand, a reflux current flows along a path of the rectifier elementD10, the W-phase, the U-phase, and the switching element SW4.

During a time from 5T/6 to T, the controller 19 turns on the switchingelements SW3 and SW5 and turns off the other switching elements. At thistime, the impedance of the switching element SW3, the W-phase, theV-phase, and the switching element SW5, from one end of the smoothingcapacitor SC to the other end of the smoothing capacitor SC is lowered.

As a result, the input impedance viewed from the smoothing capacitor SCis lowered and the rectified output V_(out) is supplied between theW-phase and the V-phase of the three-phase brushless motor M from therectifier circuit 15.

On the other hand, the impedance between one end of the smoothingcapacitor SC, and the U-phase, the V-phase, and the W-phase is raised byturning off the switching element SW3 within that time range. As aresult, the input impedance viewed from the smoothing capacitor SC israised and the supply of the rectified output V_(out) between theW-phase and the V-phase from the rectifier circuit 15 is stopped. On theother hand, a reflux current flows along a path of the rectifier elementD10, the W-phase, the V-phase, and the switching element SW5.

By combining turning ON and OFF of the switching elements SW1 to SW6, asdescribed above, the switching circuit 17 can switch between an ON-statewhich is a state in which the input impedance viewed from the smoothingcapacitor SC is low and an OFF-state which is a state in which the inputimpedance is high.

By causing the switching circuit 17 to switch between the ON-state andthe OFF-state, it is possible to switch between a state in which therectified output V_(out) is output from the rectifier circuit 15 to thethree-phase brushless motor M and a state in which an output of therectified output V_(out) to the three-phase brushless motor M stops, androtate the rotor of the three-phase brushless motor M.

A time in which the switching elements SW1 to SW3 are turned on, turnedoff, and turned on again in the switching operation of the switchingelement SW1 during the time of 0 to T/3, the switching operation of theswitching element SW2 during the time of T/3 to 2T/3, and the switchingoperation of the switching element SW3 during the time of 2T3 to T isreferred to as a “switching period T_(sw)” (see FIG. 6).

In a switching operation, a ratio of the length of a time in which eachof the switching element SW1 to SW3 is turned on to the switching periodT_(sw) is defined as a “duty ratio.” Further, a period in which each ofthe switching elements SW1 to SW3 performs a switching operation isreferred to as a “power supply period T_(ps)”.

As described above, a period in which each of the switching elements SW1to SW3 performs a switching operation, that is, the “power supply periodT_(ps)”, is a period of T/3. That is, the power supply period T_(ps) isa time of ⅓ of the rotation period of the rotor. In this way, aconduction scheme in which the power supply period T_(ps) is a time of ⅓of the rotation period of the rotor is generally referred to as a“120-degree conduction scheme”.

The controller 19 inputs a signal in which an ON signal and an OFFsignal are switched at a high speed at the switching period T_(sw)shorter than an AC period of the AC input V_(in) to the control pole ofeach of the switching elements SW1 to SW3 of the switching circuit 17during the power supply period T_(ps). Thus, the switching circuit 17can switch between the ON-state in which the input impedance viewed fromthe smoothing capacitor SC is low and the OFF-state in which the inputimpedance is high at a high speed at the switching period T_(sw) duringthe power supply period T_(ps) under the control of the controller 19.

In addition, the controller 19 determines the duty ratio of each of theswitching elements SW1 to SW3 of the switching circuit 17 which is aninverter circuit based on a predetermined condition and adjusts a ratioduring the switching period T_(sw) of the ON signal and the OFF signalinput to the control poles of the switching elements SW1 to SW3 based onthe determined duty ratio (PWM control).

As will be described below, in the embodiment, the duty ratio ismodulated in accordance with a variation of a voltage of the smoothingcapacitor SC (the rectified output V_(out)). Accordingly, hereinafter, afinally calculated duty ratio is referred to as a “modulation duty ratioDR”. A specific calculation method for the modulation duty ratio DR willbe described in detail later.

By adjusting the modulation duty ratio DR in the switching circuit 17,it is possible to adjust an output time of the rectified output V_(out)to the load LO at the switching period T_(sw). That is, it is possibleto adjust a sum of the output times of the rectified output V_(out)during the power supply period T_(ps) and adjust an average value ofcurrents and voltages supplied during the power supply period T_(ps).

In the embodiment, the controller 19 includes a measurement unit 191connected in parallel to the smoothing capacitor SC. The measurementunit 191 is a voltmeter that measures the voltage (the rectified outputV_(out)) of the smoothing capacitor SC.

The measurement unit 191 is configured as, for example, a circuit inwhich a plurality of resistive elements dividing the voltage of thesmoothing capacitor SC are connected in series.

In this case, the controller 19 connects any one of the plurality ofresistive elements connected in series to an A/D converter included inthe controller 19. Thus, the controller 19 can monitor the voltage ofthe smoothing capacitor SC.

1-4. Control Method for Driving Device in First Embodiment

Hereinafter, a control method for the driving device 100 according tothe embodiment will be described with reference to FIG. 7. FIG. 7 is aflowchart illustrating the control method for the driving device.

When the driving device 100 starts controlling the three-phase brushlessmotor M, the controller 19 measures a current voltage of the smoothingcapacitor SC from the measurement unit 191 (step S1).

As will be described below, a current voltage measurement value of thesmoothing capacitor SC is used to calculate a future modulation dutyratio DR. Accordingly, the controller 19 stores the voltage measurementvalue of the smoothing capacitor SC in association with a time at whichthe voltage measurement value is measured in a storage region of thestorage element of the controller 19.

In another embodiment, the controller 19 may store the voltagemeasurement value of the smoothing capacitor SC in association with anumber indicating at which time the voltage measurement value of thesmoothing capacitor SC is measured after start of the measurement.

Subsequently, the controller 19 calculates a difference between aninstruction value of a rotation speed of the three-phase brushless motorM set in the controller 19 or in an external device by a user and acurrent actual rotation speed of the rotor of the three-phase brushlessmotor M (step S2). The instruction value of the rotation speed of thethree-phase brushless motor M is referred to as a target rotation speed.

For example, the rotation speed of the rotor of the three-phasebrushless motor M can measure the rotation speed based on the number ofpulses per unit time input from an encoder (not illustrated) provided inan output rotation shaft of the rotor. Alternatively, for example, thecontroller 19 can measure the rotation speed of the rotor based on alength of time of being ON or OFF in the rotor detection element or alength of one period of being ON and OFF.

After the controller 19 calculates the difference between the targetrotation speed and the actual rotation speed of the rotor, thecontroller 19 calculates a duty ratio before the modulation of each ofthe switching elements SW1 to SW3 based on the difference (step S3). Theduty ratio before the modulation calculated in step S3 is setindependently from the variation in the rectified output V_(out), andthus referred to as a “basic duty ratio DR_(B)”.

In addition, in the embodiment, each of the switching elements SW1 toSW3 performs a switching operation in accordance with the 120-degreeconduction scheme. Therefore, the basic duty ratio DR_(B) of each of theswitching elements SW1 to SW3 has a different phase. The basic dutyratio DR_(B) is set independently from the rectified output V_(out).Further, the sum is a constant value independent from a time other thana variation based on a target rotation speed or the like.

For example, when the target rotation speed is greater than the actualrotation speed of the rotor, the controller 19 increases the currentlyset basic duty ratio DR_(B) and calculates the basic duty ratio DR_(B).Conversely, when the target rotation speed is less than the actualrotation speed of the rotor, the controller 19 decreases the currentlyset basic duty ratio DR_(B) and calculates the basic duty ratio DR_(B).

In step S2 described above, the user may set the power instruction valueto be output to the three-phase brushless motor M. The power instructionvalue to be output to the three-phase brushless motor M is referred toas target power. In this case, the controller 19 may calculate thedifference between the target power and an actually measured value ofthe power actually input to the three-phase brushless motor M.

The power to be input to the three-phase brushless motor M can becalculated, for example, when the controller 19 multiplies a currentvalue obtained by providing a detection mechanism (not illustrated) fora current flowing from the rectifier circuit 15 to the switching circuit17 by the voltage of the smoothing capacitor SC detected by themeasurement unit 191.

The detection mechanism for the current can be realized, for example, byproviding current detection resistors along a path from the switchingelements SW4 to SW6 to a surface lower end of the smoothing capacitorSC. The controller 19 can measure a current flowing to the switchingcircuit 17 based on a measured value of a difference between voltages ofboth ends of the resistors.

In this case, in step S3 described above, when the target power isgreater than the power actually input to the three-phase brushless motorM, the controller 19 calculates the basic duty ratio DR_(B) byincreasing the currently set basic duty ratio DR_(B). Conversely, whenthe target power is less than the power actually input to thethree-phase brushless motor M, the controller 19 calculates the basicduty ratio DR_(B) by decreasing the currently set basic duty ratioDR_(B).

After the basic duty ratio DR_(B) is calculated in this way, thecontroller 19 calculates the modulation duty ratio DR modulated inaccordance with a variation in the voltage of the smoothing capacitor SCin each of the switching elements SW1 to SW3 using the basic duty ratioDR_(B) (step S4).

In the embodiment, the controller 19 calculates the modulation dutyratio DR to be set by reading the previous voltage measurement value ofthe smoothing capacitor SC measured in step S1 a predetermined timebefore from the storage region and correcting the basic duty ratioDR_(B) based on a reciprocal of an absolute value of the voltagemeasurement value.

For example, the controller 19 calculates the modulation duty ratioDR(t) to be set at current time t by the following expression:

DR(t)={(A*V _(ave))/V(t−t′)}*DR_(B)(t)+(1−A)*DR_(B)(t)

(A: a positive constant equal to or less than 1, V_(ave): an averagevalue of voltages of the smoothing capacitor SC, V(t−t′): a voltagemeasurement value of the smoothing capacitor SC time t′ previously, andDR_(B)(t): the basic duty ratio DR_(B) at time t).

The modulation duty ratio DR(t) is further rewritten as follows:

DR(t)=[{(A*V _(ave))/V(t−t′)}+(1−A)]*DR_(B)(t)

That is, the modulation duty ratio DR(t) is a multiple of[{(A*V_(ave))/V(t−t′)}+(1−A)] which is a value obtained by varying thebasic duty ratio DR_(B)(t) in accordance with the voltage measurementvalue of the smoothing capacitor SC. Accordingly, a value expressed as[{(A*V_(ave))/V(t−t′)}+(1−A)] is called a modulation width of themodulation duty ratio DR(t).

A value of the positive constant A equal to or less than 1 and includedin the expression for calculating the modulation duty ratio DR(t) attime t can be appropriately determined in accordance with the magnitudeor the like of a ripple voltage included in the rectified outputV_(out). The average voltage V_(ave) of voltages of the smoothingcapacitor SC can be calculated, for example, by averaging the voltagemeasurement values of the smoothing capacitor SC.

The previous voltage measurement value of the smoothing capacitor SCused in the above expression may be, for example, a voltage measurementvalue a predetermined number of values previous to the current voltagemeasurement value among the voltage measurement values stored in thestorage region when the voltage measurement value is measured at each ofpredetermined times.

After the modulation duty ratio DR is calculated in this way, thecontroller 19 calculates times of the ON-state and the OFF-state of theswitching circuit 17 at the switching period T_(sw) based on thecalculated modulation duty ratio DR (step S5).

For example, the controller 19 can calculate a time at which theswitching circuit 17 enters the ON-state as T_(sw)*DR (T_(sw): aswitching period and DR: a calculated modulation duty ratio) and cancalculate a time at which the switching circuit 17 enters the OFF-stateas T_(sw)*(1−DR).

Thereafter, during the power supply period T_(ps), the controller 19repeatedly outputs an ON-signal for causing the switching circuit 17 toenter the ON-state by a time of T_(sw)*DR and outputs an OFF-signal forcausing the switching circuit 17 to enter the OFF-state by a time ofT_(sw)*(1−DR) in the switching period T_(sw) in any of the switchingelements SW1 to SW3 (step S6).

According to the time chart illustrated in FIG. 6, a switching elementto which the signal is output can be determined among the switchingelements SW1 to SW3, for example, by confirming the ON-state and theOFF-state of the rotor detection element.

By performing steps S5 and S6 described above, the controller 19 cancontrol switching between the ON-state and the OFF-state of theswitching circuit 17 by adjusting a length in which the ON-state ismaintained and a length in which the OFF-state is maintained inaccordance with the set modulation duty ratio DR.

For example, the controller 19 receives an instruction to stop anoperation of the driving device 100 or detects abnormality of thedriving device 100 and repeatedly performs steps S1 to S6 as long as itis determined that the control of the driving device 100 does not end(as long as “No” in step S7). That is, the control of the driving device100 continues.

Conversely, when it is determined that the control of the driving device100 ends (the case of “Yes” in step S7), the controller 19 stops thecontrol of the driving device 100 after appropriately performing endsequence control as necessary.

By repeatedly performing steps S1 to S6 described above, the modulationduty ratio DR set in the controller 19 varies, as illustrated in FIG. 8.FIG. 8 is a diagram illustrating an example of a variation in amodulation duty ratio set in the driving device.

In FIG. 8, the rectified output V_(out) is indicated by a dotted line.On the other hand, the modulation duty ratio DR set at each time isindicated by white triangles and a solid line. FIG. 8 illustrates a setvalue of the modulation duty ratio DR corresponding to almost one periodof a variation in the rectified output V_(out), and the variation in theset value of the modulation duty ratio DR illustrated in FIG. 8continues during an operation of the driving device 100.

As illustrated in FIG. 8, the modulation duty ratio DR calculated inthis way varies periodically between a first duty ratio DR1 and a secondduty ratio DR2 greater than the first duty ratio DR1 to correspond to aperiodic voltage variation of the rectified output V_(out). Themagnitudes of the first duty ratio DR1 and the second duty ratio DR2 canbe determined in accordance with, for example, the constant A in theforegoing expression for calculating the duty ratio. The second dutyratio DR2 may be a maximum duty ratio which can be set in the powersupply device 1.

Here, a direction of the variation in the modulation duty ratio DR isopposite to that of the variation in the voltage of the rectified outputV_(out). This is because the modulation duty ratio DR in the embodimentis calculated based on the reciprocal of the voltage measurement valueof the smoothing capacitor SC.

The periodic variation in the modulation duty ratio DR deviates from theperiodic variation in the rectified output V_(out) by a predeterminedtime. Specifically, a timing t2 at which the modulation duty ratio DRbecomes the second duty ratio DR2 is delayed by a time t2 from a timingt1 at which the rectified output V_(out) becomes a minimum value V_(min)in FIG. 8.

A timing t4 at which the modulation duty ratio DR becomes the first dutyratio DR1 is also delayed by the time t′ from a timing t3 at which therectified output V_(out) becomes a maximum value V_(rmax).

A delay width (time t′) of the variation in the modulation duty ratio DRwith respect to the variation in the rectified output V_(out) can bedetermined in accordance with, for example, V(t−t′) in the foregoingexpression for calculating the modulation duty ratio DR. That is, theduty ratio DR can be determined in accordance with a certain previousvoltage measurement value used to calculate the modulation duty ratio DRamong the voltage measurement values of the smoothing capacitor SC.

The rectified output V_(out) varies to correspond to the frequency ofthe AC input V_(in). Specifically, for example, when the frequency ofthe AC input V_(in) increases, the period of the variation in therectified output V_(out) is also shortened. Accordingly, the controller19 may determine the time t′ which is a specific deviation width betweenthe variation in the modulation duty ratio DR and the variation in therectified output V_(out) based on an AC period of the AC input V_(in).

For example, when the AC period of the AC input V_(in) is shortened, thecontroller 19 selects a voltage measurement value measured at a timecloser to the present than a previous voltage measurement value of thesmoothing capacitor SC selected before a decrease in the AC period andcalculates the modulation duty ratio DR. That is, the time t′ in theforegoing expression is set to be small.

Thus, even when the AC period of the AC input V_(in) varies, thedeviation in the variation in the modulation duty ratio DR with respectto the variation of the rectified output V_(out) can be maintained as anoptimum value. Specifically, for example, it is possible to constantlymaintain the deviation between a phase of the variation in the rectifiedoutput V_(out) and a phase of the variation in the modulation duty ratioDR.

When the timing at which the modulation duty ratio DR becomes the secondduty ratio DR2 deviates by the predetermined time t′ from the timing atwhich the voltage of the rectified output V_(out) becomes the minimumvalue V_(rmin), as illustrated in FIG. 8, the timing at which themodulation duty ratio DR becomes the second duty ratio DR2, that is, atiming at which the modulation width of the modulation duty ratio DRbecomes a maximum value, is included in a period in which the voltage ofthe rectified output V_(out) increases from the minimum value V_(rmin)to the maximum value V_(rmax).

By including the timing at which the modulation duty ratio DR becomesthe second duty ratio DR2 in the period in which the voltage of therectified output V_(out) increases from the minimum value V_(rmin) tothe maximum value V_(rmax), it is possible to maintain a state in whichthe modulation duty ratio DR is large within the period.

Within the period in which the voltage of the rectified output V_(out)increases from the minimum value V_(rmin) to the maximum value V_(rmax),an input current with which the smoothing capacitor SC is charged flowsin the smoothing capacitor SC.

Accordingly, by including the timing at which the modulation duty ratioDR becomes the second duty ratio DR2 in the period in which the voltageof the rectified output V_(out) increases from the minimum valueV_(rmin) to the maximum value V_(rmax), that is, a period from a time atwhich the input current is generated to a time at which the voltage ofthe smoothing capacitor SC becomes the maximum value V_(rmax) andincreasing the modulation duty ratio DR during the charging period ofthe smoothing capacitor SC, it is possible to supply more power from therectifier circuit 15 to the load LO.

On the other hand, in a period other than the period in which thevoltage of the rectified output V_(out) increases from the minimum valueV_(rmin) to the maximum value V_(rmax), in particular, in a period inwhich the voltage of the rectified output V_(out) decreases from themaximum value V_(rmax), the modulation duty ratio DR is set to arelatively small set value.

The decrease in the voltage of the rectified output V_(out) in theperiod means that power is supplied to the load LO by discharging thesmoothing capacitor SC. Accordingly, by setting the modulation dutyratio DR in the period in which the voltage of the rectified outputV_(out) decreases to a small set value, it is possible to reduce a powersupply amount to the load LO in the period and reduce a discharge amountof the smoothing capacitor SC.

By reducing the discharge amount of the smoothing capacitor SC, it ispossible to reduce a charge amount for setting the voltage of thesmoothing capacitor SC to the maximum value V_(rmax).

By reducing the discharge amount and the charge amount of the smoothingcapacitor SC in this way, it is possible to reduce outflow and inflow ofa current of the smoothing capacitor SC and suppress deterioration ofthe smoothing capacitor SC over time. When an electrolytic capacitor isused as the smoothing capacitor SC, a lifespan of the electrolyticcapacitor is considerably affected by temperature, but the electrolyticcapacitor has relatively large internal resistance. Therefore, when acurrent flowing in and out of the smoothing capacitor SC, a so-calledripple current, is large, heating is caused inside the smoothingcapacitor SC and the lifespan is reduced.

In the embodiment, since the ripple current can be reduced,deterioration of the smoothing capacitor SC due to the heating can besuppressed and reliability of the circuit can be improved. Since anelectrolytic capacitor that has capacitance less as the ripple currentis smaller can be used, it is possible to reduce manufacturing cost ofthe circuit.

Additionally, as described above, even when an amount of power varyingover time is supplied to the load LO during driving of the load LO, inparticular, when the load LO has a large inertia moment such as in therotor of the three-phase brushless motor M or reactivity to supply powerto the load LO is bad, pulsation does not occur in the driving of theload LO and the load LO can be stably driven in accordance with anaverage value of a given power amount.

1-5. Experiment Results

Hereinafter, experiment results for verifying an effect of delaying thevariation in the modulation duty ratio DR by a predetermined time withrespect to the variation in the voltage of the smoothing capacitor SCwill be described.

To verify the effect, a load current flowing in the U-phase of thethree-phase brushless motor was measured when the delay width (time t′)of the timing at which the modulation duty ratio DR becomes the secondduty ratio DR2 with respect to the timing at which the voltage of thesmoothing capacitor SC becomes the minimum value V_(rmin) was setvariously and the driving device 100 was operated.

First, how the load current changes depending on whether a variation inthe modulation duty ratio DR is delayed with respect to a variation inthe voltage of the smoothing capacitor SC will be described withreference to FIG. 9.

In the lower drawing of FIG. 9, a plot of a solid line indicates anabsolute value of a load current when there is a delay (time t′=t2′). Onthe other hand, a plot of a dotted line indicates an absolute value of aload current when there is no delay (time t′=0).

As illustrated in FIG. 9, by delaying the variation in the modulationduty ratio DR with respect to the variation in the voltage of thesmoothing capacitor SC within a period in which the voltage of thesmoothing capacitor SC increases from the minimum value V_(rmin) to themaximum value V_(rmax), a larger current flows in the U-phase of thethree-phase brushless motor than when the variation in the modulationduty ratio DR is not delayed. That is, by delaying the variation in theduty ratio with respect to the variation in the voltage of the smoothingcapacitor SC, a current can flow to the U-phase of the three-phasebrushless motor more actively while the smoothing capacitor SC ischarged.

On the other hand, by delaying the variation in the modulation dutyratio DR with respect to the variation in the voltage of the smoothingcapacitor SC within a period in which the voltage of the smoothingcapacitor SC decreases, a smaller current flows in the U-phase of thethree-phase brushless motor than when the variation in the modulationduty ratio DR is not delayed. That is, by delaying the variation in theduty ratio with respect to the variation in the voltage of the smoothingcapacitor SC, the current flowing in the U-phase of a three-shapebrushless motor is reduced during discharging of the smoothing capacitorSC.

Next, how a variation width of the voltage of the smoothing capacitor SC(a ripple voltage) and a current flowing in and out of the smoothingcapacitor SC (a charging current and a discharging current: a so-calledripple current) are changed at the time of variously changing the timet′ by which the variation in the modulation duty ratio DR is delayedwith respect to the variation in the voltage of the smoothing capacitorSC will be described with reference to FIG. 10.

In a verification result illustrated in FIG. 10, the time t′ by whichthe variation in the modulation duty ratio DR is delayed with respect tothe variation in the voltage of the smoothing capacitor SC was set to 0(no delay), t1′, t2′, t3′, and t4′ (where t1′<t2′<t3′<t4′). The ripplevoltage was set to a difference between the maximum value V_(rmax) andthe minimum value V_(rmin) of the rectified output V_(out). Further, theripple current was set to a root mean square (RMS) value of currentsflowing in and out of the smoothing capacitor SC.

As illustrated in FIG. 10, by delaying the variation in the modulationduty ratio DR with respect to the variation in the voltage of thesmoothing capacitor SC, both the ripple voltage and the ripple currentare considerably decreased more than when there is no delay. That is, bydelaying the variation in the modulation duty ratio DR with respect tothe variation in the voltage of the smoothing capacitor SC, a currentamount flowing in and out of the smoothing capacitor SC is considerablyreduced.

In addition, by delaying the variation in the modulation duty ratio DRwith respect to the variation in the voltage of the smoothing capacitorSC, the ripple voltage is decreased and the rectified output V_(out) ofwhich a voltage is more stable and a ripple ratio is less is output.

Further, as illustrated in FIG. 10, the ripple voltage and the ripplecurrent are more decreased as the length of the time t′ by which thevariation in the modulation duty ratio DR is delayed with respect to thevariation in the voltage of the smoothing capacitor SC is larger. Bydecreasing the ripple current, it is possible to obtain the effect ofimproving reliability of the circuit by suppressing deterioration of thesmoothing capacitor SC or the effect of reducing cost by using acapacitor with lower capacitance as the smoothing capacitor SC.

Here, it is preferable not to excessively increase the time t′ by whichthe variation in the modulation duty ratio DR is delayed with respect tothe variation in the voltage of the smoothing capacitor SC. In a resultobtained by setting excessively long time t′ by which the variation inthe modulation duty ratio DR is delayed, for example, as illustrated inFIG. 11, the timing t2 at which the modulation duty ratio DR becomes thesecond duty ratio DR2 is assumed to arrive at a time near the timing t3at which the voltage of the smoothing capacitor SC becomes the maximumvalue V_(rmax). In the example illustrated in FIG. 11, t5′ is set as thetime t′ by which the variation in the modulation duty ratio DR isdelayed with respect to the variation in the voltage of the smoothingcapacitor SC.

In the foregoing case, as illustrated in FIG. 11, a load current to theload LO indicated by a solid line in the lower drawing of FIG. 11 isexcessive, in particular, in a time range in which the voltage of thesmoothing capacitor SC is near the maximum value V_(rmax). A plotindicated by a dotted line in the lower drawing of FIG. 11 is anabsolute value of a load current when t2′ is set as the time t′ by whichthe variation in the modulation duty ratio DR is delayed with respect tothe variation in the voltage of the smoothing capacitor SC.

This is because the modulation duty ratio DR increases when the voltageof the smoothing capacitor SC is near the maximum value V_(rmax) in acase where the timing t2 at which the modulation duty ratio DR becomesthe second duty ratio DR2 is near the timing t3 at which the voltage ofthe smoothing capacitor SC is the maximum V_(rmax). That is, this isbecause a current is actively supplied to the load LO when the voltageof the rectified output V_(out) is large.

When an excessive current flows in the load LO, a conduction loss ineach element or the like included in the rectifier circuit 15 and theswitching circuit 17 is large. As a result, energy efficiency degradesand/or the power supply device 1 overheats due to the large conductionloss. As a result, the power supply device 1 operates abnormally orbreaks down.

When an excessive current flows in the load LO, for example, it isnecessary to use switching elements that have large current capacity andoperate at a high speed as the switching elements SW1 to SW6 of theswitching circuit 17, and thus high cost of the driving device 100 maybe caused.

Second Embodiment 2-1. Overview of Second Embodiment

In the foregoing first embodiment, the basic duty ratio DR_(B) of eachof the switching elements SW1 to SW3 is a constant value independentfrom a time other than the variation in accordance with the targetrotation speed, the target power, or the like. The present invention isnot limited thereto and the basic duty ratio DR_(B) of each of theswitching elements SW1 to SW3 may also vary over time other than thevariation in accordance with the target rotation speed, the targetpower, or the like. That is, the basic duty ratio DR_(B) may beexpressed as a function of a time.

2-2. Example 1

Hereinafter, Example 1 in which the basic duty ratio DR_(B) of each ofthe switching elements SW1 to SW6 is set as a sinusoidal waveformvarying at a predetermined period with a different phase from that ofthe basic duty ratio DR_(B) of the other switching elements SW1 to SW6will be described with reference to FIG. 12 to FIG. 15. FIG. 12 is adiagram illustrating an example of a basic duty ratio set in eachswitching element. FIG. 13 is a diagram illustrating an example of amodulation duty ratio of each switching element. FIG. 14 is a diagramillustrating an example of a modulation duty ratio when a variationfrequency of the basic duty ratio is more increased. FIG. 15 is adiagram illustrating an example of a modulation duty ratio when avariation frequency of the basic duty ratio is further more increased.

In FIG. 12 to FIG. 15, the basic duty ratio DR_(B) and the modulationduty ratio DR indicated by a thick solid line are a U-phase duty ratio.The basic duty ratio DR_(B) and the modulation duty ratio DR indicatedby a thick dotted line are a V-phase duty ratio. The basic duty ratioDR_(B) and the modulation duty ratio DR indicated by a thick one-dotchain line are a W-phase duty ratio. In the graphs of FIG. 12 to FIG.15, a modulation width is indicated by a thick solid line and a value ofthe rectified output V_(out) is indicated by a thick dotted line.

In Example 1, a high side of the switching circuit 17, that is, the dutyratios of the switching elements SW1 to SW3, varies in a sinusoidalwaveform. The switching elements SW4 to SW6 on the low side vary theduty ratios complementarily with the corresponding switching elementsSW1 to SW3. In the following drawings, the duty ratio of the switchingelement SW1 is referred to as a “U-phase”, the duty ratio of theswitching element SW2 is referred to as a “V-phase”, and the duty ratioof the switching element SW3 is referred to as a “W-phase”.

As illustrated in FIG. 12, in Example 1, when the U-phase basic dutyratio DR_(B) varies in a sinusoidal waveform and the U-phase serves as areference, the V-phase basic duty ratio DR_(B) varies in a sinusoidalwaveform with a phase delayed by 120 degrees and the W-phase basic dutyratio DR_(B) varies in a sinusoidal waveform with a phase delayed by 240degrees. That is, the basic duty ratio DR_(B) of each phase has amutually different phase.

Accordingly, the basic duty ratio DR_(B) of each phase is expressed asthe following expression. In the following expressions, C is anamplitude of the basic duty ratio which varies in accordance with atarget rotation speed and/or target power within a range of 0≤C≤1/2 andis a value independent from a time and the rectified output V_(out).When C=0, the basic duty ratio of each phase is 1/2 together. ω is anangular velocity and is expressed as 2πf using a frequency f.

In addition, t is a time,

-   -   the U-phase basic duty ratio: 1/2+C*sin(ωt)    -   the V-phase basic duty ratio: 1/2+C*sin(ωt−2π/3)    -   the W-phase basic duty ratio: 1/2+C*sin(ωt−4π/3)

By setting the basic duty ratio DR_(B) varying in the sinusoidalwaveform in this way, the switching circuit 17 can output a voltage anda current of the sinusoidal waveform that has a frequency determined atω to the output unit 13.

In the embodiment, the controller 19 calculates the modulation dutyratio DR by multiplying the amplitude C of the basic duty ratio DR_(B)of the sinusoidal waveform expressed above by the modulation width[{(A*V_(ave))/V(t−t′)}+(1−A)] varying in accordance with the voltagemeasurement value of the smoothing capacitor SC, as described in thefirst embodiment. Accordingly, in Example 1, the modulation duty ratioDR of each phase is expressed as follows:

-   -   the U-phase modulation duty ratio:        1/2+[{(A*V_(ave))/V(t−t′)}+(1−A)]*{C*sin(ωt)}    -   the V-phase modulation duty ratio:        1/2+[{(A*V_(ave))/V(t−t′)}+(1−A)]*{C*sin(ωt−2π/3)}    -   the W-phase modulation duty ratio:        1/2+[{(A*V_(ave))/V(t−t′)}+(1−A)]*{C*sin(ωt−4π/3)}

For the modulation duty ratio DR of each phase expressed above, aconsiderable variation is not seen in the period of the variation, asillustrated in FIG. 13. On the other hand, in the modulation duty ratioDR, the amplitude of the sinusoidal waveform included in the basic dutyratio DB is modulated by C*[{(A*V_(ave))/V(t−t′)}+(1−A)]. Therefore, thewaveform of the modulation duty ratio DR has a waveform slightlydistorted from the sinusoidal waveform. Here, the modulation width ofthe amplitude of the sinusoidal waveform of the modulation duty ratio DRbecomes the largest within a period until the rectified output V_(out)becomes a maximum as in the first embodiment.

The method of setting the modulation duty ratio DR can be applied evenwhen the frequency of the basic duty ratio DR_(B) is different from thecase illustrated in FIG. 11. Specifically, as illustrated in FIG. 14 andFIG. 15, even when the angular velocity ω of the basic duty ratio DR_(B)is set to be large, the modulation width of the amplitude of thesinusoidal waveform portion of the modulation duty ratio DR of eachphase is the largest within a period until the rectified output V_(out)becomes a maximum as in the modulation duty ratio DR of the firstembodiment. That is, the modulation duty ratio DR can be appropriatelyset for any angular velocity ω of the basic duty ratio DR_(B), that is,any frequency.

Further, the modulation of the basic duty ratio D_(B) can also beapplied similarly to the basic duty ratio D_(B) with any shape otherthan a sinusoidal waveform varying at a predetermined period, forexample, a trapezoidal basic duty ratio.

2-3. Example 2

Hereinafter, Example 2 in a case in which the basic duty ratio DR_(B) ofeach of the switching elements SW1 to SW6 is modulated by a two-phasemodulation scheme will be described with reference to FIG. 16 to FIG.19. FIG. 16 is a diagram illustrating another example of the basic dutyratio set in each switching element. FIG. 17 is a diagram illustrating amodulation width of a modulation duty ratio and another example of themodulation duty ratio of each switching element. FIG. 18 is a diagramillustrating a modulation width of the modulation duty ratio and anotherexample of the modulation duty ratio when the variation frequency of thebasic duty ratio is more increased. FIG. 19 is a diagram illustrating amodulation width of the modulation duty ratio and another example of themodulation duty ratio when the variation frequency of the basic dutyratio is further more increased.

In FIG. 16 to FIG. 19, the basic duty ratio DR_(B) and the modulationduty ratio DR indicated by a thick solid line are a U-phase duty ratio.The basic duty ratio DR_(B) and the modulation duty ratio DR indicatedby a thick dotted line are a V-phase duty ratio. The basic duty ratioDR_(B) and the modulation duty ratio DR indicated by a thick one-dotchain line are a W-phase duty ratio. In the graphs of FIG. 17 to FIG.19, a modulation width is indicated by a thick solid line and a value ofthe rectified output V_(out) is indicated by a thick dotted line.

The two-phase modulation scheme is a modulation scheme of a duty ratioin which two switching elements SW1 and SW2 are subjected to the PWMmodulation among three switching elements SW1 to SW3 and the remainingone of the switching elements SW1 to SW3 is in the OFF-state. Thetwo-phase modulation scheme has the advantage of decreasing a switchingdamage compared to Example 1 since a period in which one of theswitching elements SW1 to SW3 is in the OFF-state is provided.

A temporal variation of the basic duty ratio DR_(B) set at each phase isillustrated in FIG. 16. The basic duty ratio DR_(B) is expressed as thefollowing expression. The following expression represents the U-phasebasic duty ratio DR_(B). The V-phase and W-phase basic duty ratiosDR_(B) delay the U-phase basic duty ratio DR_(B) by 120 degrees and 240degrees, respectively.

In the following expressions, D is an amplitude of the basic duty ratiowhich varies in accordance with a target rotation speed, target power,or the like and is a value independent from a time and the rectifiedoutput V_(out). ω is an angular velocity. In addition, n is an integer.In addition, t is a time,

-   -   (i) D*sin(ωt)(2πn≤ωt<2πn+2π/3)    -   (ii) D*sin(ωt−π/3)(2πn+2π/3≤ωt<2πn+4π/3)    -   (iii) 0(2πn+4π/3ωt<2π(n+1))

In the embodiment, the controller 19 calculates the modulation dutyratio DR by multiplying the amplitude D of the foregoing basic dutyratio DR_(B) by the modulation width [{(A*V_(ave))/V(t−t′)}+(1−A)]varying in accordance with the voltage measurement value of thesmoothing capacitor SC, as described in the first embodiment. Themodulation duty ratio DR of each phase calculated in this way varies inaccordance with the foregoing variation width as in Example 1 describedabove, as illustrated in FIG. 17 to FIG. 19.

As illustrated in FIG. 17 to FIG. 19, the method of setting themodulation duty ratio DR of the embodiment can be applied to anyfrequency of the basic duty ratio DR_(B), as in Example 1 describedabove, even when the basic duty ratio DR_(B) modulated by the two-phasemodulation scheme is used.

Even when the basic duty ratio DR_(B) is arbitrarily varied as afunction of a time, as described above in Examples 1 and 2, by furtherlengthening a state in which the input impedance viewed from thesmoothing capacitor SC is low within a period from a time at which aninput current input to the smoothing capacitor is generated to a time atwhich the voltage of the smoothing capacitor SC becomes a maximum, thatis, within a period in which the smoothing capacitor SC is charged, itis possible to reduce a current amount flowing in the smoothingcapacitor SC and suppress deterioration of the smoothing capacitor SC.

Third Embodiment 3-1. Overview of Third Embodiment

In the foregoing first embodiment, a current amount flowing in and outof the smoothing capacitor SC is reduced by setting a duty ratio whichis a ratio to the switching period T_(sw) during a time in which theswitching circuit 17 is in the ON-state to a large set value within thecharging period of the smoothing capacitor SC, that is, within theperiod in which the voltage of the smoothing capacitor SC increases fromthe minimum value V_(rmin) to the maximum value V_(rmax), and bysupplying power to the load LO actively within the charging period ofthe smoothing capacitor SC.

When a brushless motor such as the three-phase brushless motor M isparticularly used as the load LO, so-called advance angle control inwhich a start timing of the power supply period T_(ps) which is a periodin which power is supplied to the load LO is adjusted for efficientdriving of the motor is performed.

In the advance angle control, a phase of a current flowing in a statorwhich is an electromagnet is adjusted with respect to rotation of arotor of a brushless motor. The phase of the current flowing in theelectromagnet of the stator is delayed with respect to the phase of thevoltage applied to a coil of the stator mainly by an inductancecomponent that the coil of the stator has. The delay is considerableparticularly when a current flowing in the coil of the stator isincreased and/or a period of the current is shortened in order to rotatethe rotor at a high speed.

Therefore, as illustrated in FIG. 6, even when a voltage is applied tothe U-phase, the V-phase, and the W-phase of the three-phase brushlessmotor M at a timing at which the rotor is detected in a rotor detectionelement, a timing at which a current starts to flow in the U-phase, theV-phase, and the W-phase is later than a timing at which the rotor isdetected in the rotor detection element. Thus, a phase of a rotatingmagnetic field from the stator generated by the current flowing in theU-phase, the V-phase, and the W-phase is not an optimum phase forrotation of the rotor. As a result, for example, a rotation torque ofthe rotor is lowered, and thus the rotor may not be rotated efficiently.

Accordingly, in the third embodiment, a start timing of the power supplyperiod T_(ps), that is, a timing at which an output of the voltage ofthe rectified output V_(out) starts, is set to be earlier than a timingat which the rotor is detected by a rotor detection element, asillustrated in FIG. 20. Thus, it is possible to start supplying acurrent to a coil of a stator at a timing at which the rotor is detectedby the rotor detection element or a timing close to the detected timing.

Hereinafter, a magnitude when a start timing of the power supply periodT_(ps) is advanced temporally before is referred to as an “advance angleδ”. The advance angle δ that is 0 is assumed to mean that the starttiming of the power supply period T_(ps) is identical with a timing atwhich the rotor is detected by the rotor detection element.

In the third embodiment, further, to actively supply power to thethree-phase brushless motor M which is the load LO within the chargingperiod of the smoothing capacitor SC, that is, within the period inwhich the voltage of the smoothing capacitor SC increases from theminimum value V_(rmin) to the maximum value V_(rmax), a start timing ofthe power supply period T_(ps) is set to a timing earlier than a timingadjusted to efficiently rotate rotor within the period. That is, theadvance angle δ within the period is set to be greater than the advanceangle δ for efficiently rotating the rotor.

For example, when a rotation speed of the rotor is not large and a phaseof a current of the coil of the stator is not considerably delayed froma phase of an application voltage despite the advance angle δ of 0and/or there is no considerable influence on rotation efficiency of therotor despite deviation of the phase of the current of the coil of thestator from the phase of the application voltage, the advance angle δmay be 0.

In the third embodiment, only a method of actively supplying power (acurrent) to the load LO within the charging period of the smoothingcapacitor SC differs from that of the first embodiment, and theconfiguration of the driving device 100, a function of each constituentelement, and the like are the same as those of the driving device 100according to the first embodiment.

Accordingly, hereinafter, only a control method for the driving device100 according to the third embodiment will be described and descriptionof the configuration and the like of the driving device 100 will beomitted.

3-2. Control Method for Driving Device According to Third Embodiment

Hereinafter, a control method for the driving device 100 according to athird embodiment will be described with reference to FIG. 21. FIG. 21 isa flowchart illustrating the control method for the driving deviceaccording to the third embodiment.

When the driving device 100 starts controlling the three-phase brushlessmotor M, the controller 19 measures a current voltage of the smoothingcapacitor SC from the measurement unit 191 (step S1′).

Subsequently, the controller 19 calculates a difference between aninstruction value of the rotation speed of the three-phase brushlessmotor M set in the controller 19 or in an external device by a user anda current actual rotation speed of the rotor of the three-phasebrushless motor M (step S2′). The instruction value of the rotationspeed of the three-phase brushless motor M is referred to as a targetrotation speed.

After the controller 19 calculates the difference between the targetrotation speed and the actual rotation speed of the rotor, thecontroller 19 calculates a basic duty ratio DR_(B) based on thedifference (step S3′).

For example, when the target rotation speed is greater than the actualrotation speed of the rotor, the controller 19 increases the currentlyset basic duty ratio DR_(B) and calculates a new basic duty ratioDR_(B). Conversely, when the target rotation speed is less than theactual rotation speed of the rotor, the controller 19 decreases thecurrently set basic duty ratio DR_(B) and calculates a new basic dutyratio DR_(B).

In step S2′ described above, the user may set the target power. In thiscase, the controller 19 may calculate the difference between the targetpower and an actually measured value of the power actually input to thethree-phase brushless motor M.

In this case, in step S3′ described above, when the target power isgreater than the power actually input to the three-phase brushless motorM, the controller 19 calculates the new basic duty ratio DR_(B) byincreasing the currently set basic duty ratio DR_(B). Conversely, whenthe target power is less than the power actually input to thethree-phase brushless motor M, the controller 19 calculates the newbasic duty ratio DR_(B) by decreasing the currently set basic duty ratioDR_(B).

After the basic duty ratio DR_(B) is calculated, the controller 19calculates the advance angle δ varying in accordance with the variationin the voltage of the smoothing capacitor SC (step S4′). In theembodiment, specifically, for example, the advance angle δ is calculatedin accordance with the flowchart illustrated in FIG. 22.

First, the controller 19 calculates an advance angle for rotating therotor of the three-phase brushless motor M efficiently (step S41′). Theadvance angle for rotating the rotor of the three-phase brushless motorM efficiently is referred to as a basic advance angle δ_(b).

Specifically, for example, the basic advance angle δ_(b) is calculatedbased on an actually measured value of a rotation speed of the rotormeasured when the basic duty ratio DR_(B) is calculated.

More specifically, for example, with reference to a table in which therotation speed of the rotor is associated with an basic advance angleδ_(b) optimum for the rotation speed of the rotor, the controller 19 cancalculate the basic advance angle δ_(b) based on an actually measuredvalue of a rotation speed of the rotor by searching for the basicadvance angle δ_(b) with which the actually measured value of therotation speed of the rotor is associated in the table.

Additionally, for example, the controller 19 can also calculate thebasic advance angle δ_(b) based on the actually measured value of therotation speed of the rotor by substituting the actually measured valueof the rotation speed of the rotor into an expression in which the basicadvance angle δ_(b) is represented as a function of the rotation speedof the rotor.

Subsequently, the controller 19 determines whether the newly calculatedbasic advance angle δ_(b) is corrected when the advance angle δ to becurrently set is calculated. Specifically, the controller 19 reads aprevious voltage measurement value of the smoothing capacitor SC apredetermined time before from the storage region and determines whetherthe voltage measurement value is less than the average voltage V_(ave)of the voltage of the smoothing capacitor SC (step S42′).

When the previous voltage measurement voltage of the smoothing capacitorSC the predetermined time before is equal to or greater than the averagevoltage V_(ave) of the voltage of the smoothing capacitor SC (“No” instep S42′), the controller 19 sets the basic advance angle δ_(b)calculated in step S41′ to the advance angle δ to be currently set (stepS43′).

Conversely, when the previous voltage measurement value of the smoothingcapacitor SC the predetermined time before is less than the averagevoltage V_(ave) of the voltage of the smoothing capacitor SC (“Yes” instep S42′), the controller 19 corrects the basic advance angle δ_(b)calculated in step S41′ based on a reciprocal of the previous voltagemeasurement value of the smoothing capacitor SC before the predeterminedtime and sets the corrected basic advance angle to the advance angle δto be currently set (step S44′).

Specifically, for example, the controller 19 calculates the advanceangle δ (6(t)) to be currently set using the following expression:

δ(t)=δ_(b)(t)*{B*V _(ave) /V(t−t′)−E}

(B and E: a positive constant (where B≥E+1), V_(ave): an average valueof voltages of the smoothing capacitor SC (voltages of the rectifiedoutput V_(out)), V(t−t′): the voltage measurement value of the smoothingcapacitor SC time t′ previously, and δ_(b)(t): a basic advance angle attime t).

The values of the positive constant B and E can be appropriatelydetermined in accordance with the magnitude or the like of the ripplevoltage included in the rectified output V_(out) when the foregoingcondition, for example, B≥E+1, is satisfied.

For example, the previous voltage measurement value of the smoothingcapacitor SC used in the foregoing expression may be a voltagemeasurement value previous by a predetermined number from the currentvoltage measurement value among the voltage measurement values stored inthe storage region when the voltage measurement value is measured foreach of predetermined times.

After the advance angle δ to be set in this way is calculated, thecontroller 19 determines the start timing of the power supply periodT_(ps) based on the calculated advance angle δ and starts the powersupply period T_(ps) at the determined start timing (step S5′).

When the power supply period T_(ps) starts, the controller 19 firstcalculates times of the ON-state and the OFF-state of a switchingoperation performed within the power supply period T_(ps) based on thebasic duty ratio DR_(B) calculated in steps S1′ to S3′ described above.

For example, the controller 19 can calculate a time in which theswitching circuit 17 is in the ON-state as T_(sw)*DR_(B) using theswitching period T_(sw) and calculate a time in which the switchingcircuit 17 is in the OFF-state as T_(sw)*(1−DR_(B)).

Subsequently, the controller 19 specifically determines a temporaltiming at which the power supply period T_(ps) starts based on thecalculated advance angle δ. Specifically, for example, the specifictiming can be calculated as follows. In the following description, thecontroller 19 counts a time from a timing at which an output of therotor detection element of a certain phase rises or falls and thedetection result is changed to a timing at which the output of the rotordetection element of another phase rises or falls and the detectionresult is changed.

As illustrated in FIG. 20, a shortest time from a timing at which thedetection result of the rotor detection element of a certain phase ischanged to a timing at which the detection result of the rotor detectionelement of another phase is changed is constant at T/6. This means thatthe phase of the rotor is changed by a constant π/3 from the change inthe detection result of the rotor detection element of a certain phaseto the change in the detection result of the rotor detection element ofanother phase. Here, a time at which the phase of the rotor is changedby π/3 is defined as T_(a).

Accordingly, for example, when the start timing of the power supplyperiod T_(ps) of the switching element SW1 is determined, the controller19 starts counting the time after the signal of the rotor detectionelement of the V-phase is turned off, and starts the power supply periodT_(ps) of the switching element SW1 at a timing at which the time iscounted as {1−δ/(π/3)}*T_(a).

That is, the controller 19 starts the power supply period T_(ps) of theswitching element SW1 at a timing the phase of the rotor is changed byπ/3−δ after the signal of the rotor detection element of the V-phase isturned off.

At the start timing of the power supply period T_(ps) of another phase,the calculation can also be performed in this way.

The time T_(a) at which the phase of the rotor is changed by π/3 ischanged in accordance with the rotation speed of the rotor. Accordingly,when the start timing of the power supply period T_(ps) of the switchingelement SW1 is determined, the controller 19 counts a time from a timeat which the signal of the rotor detection element of the W-phase isturned on to a time at which the signal of the rotor detection elementof the V-phase is turned off as the time T_(a) in advance, for example,immediately before the start timing is determined.

Conversely, the controller 19 can calculate an angular velocity of therotor as (π/3)/T_(a)=π/(3T_(a)) by counting the shortest time T_(a) froma timing at which a detection of the rotor detection element of acertain phase to a timing at which the detection result of the rotordetection element of another phase.

Thereafter, the controller 19 repeatedly performs starting power supplyperiod T_(ps) at the start timing determined in this way in any one ofthe switching elements SW1 to SW3, outputting the ON-signal for turningon the switching circuit 17 by T_(sw)*DR_(inst) at the switching periodT_(sw) during the power supply period T_(ps), and outputting theOFF-signal for turning off the switching circuit 17 byT_(sw)*(1−DR_(inst)).

By performing step S5′ described above, the controller 19 can startswitching between the ON-state and the OFF-state of the switchingcircuit 17 at the set start timing and continue the switching during thepower supply period.

For example, the controller 19 receives an instruction to stop anoperation of the driving device 100 or detects abnormality of thedriving device 100 and repeatedly performs steps S1′ to S5′ as long asit is determined that the control of the driving device 100 does not end(as long as “No” in step S6′). That is, the control of the drivingdevice 100 continues.

Conversely, when it is determined that the control of the driving device100 ends (the case of “Yes” in step S6′), the controller 19 stops thecontrol of the driving device 100.

By repeatedly performing step S1′ to S6′ described above, the advanceangle δ set in the controller 19 varies, for example, as illustrated inFIG. 23. In FIG. 23, the rectified output V_(out) is indicated by adotted line. On the other hand, the advance angle δ set at each time isindicated by white triangles and a solid line.

As illustrated in FIG. 23, the advance angle δ calculated in this wayvaries periodically between a first advance angle δ1 (an example of afirst timing) and a second advance angle δ2 (an example of a secondtiming) greater than the first advance angle δ1 to correspond to aperiodic voltage variation of the rectified output V_(out). Here, thefirst advance angle δ1 is the same as the basic advance angle δ_(b).

The second advance angle δ2 may be a maximum value of an advance anglewhich can be set in the driving device 100.

Here, a direction of a variation in the advance angle δ is opposite tothat of a variation in the voltage of the rectified output V_(out). Thisis because the advance angle δ is calculated based on a reciprocal ofthe voltage measurement value of the smoothing capacitor SC in theembodiment.

Additionally, a periodic variation in the advance angle δ deviates by apredetermined time from a periodic variation of the rectified outputV_(out). Specifically, a timing t2″ at which the advance angle δ becomesthe second advance angle δ2 is delayed by a time t′ from a timing t1″ atwhich the rectified output V_(out) becomes a minimum value V_(rmin) inFIG. 23.

A delay width of the variation in the advance angle δ with respect tothe variation in the rectified output V_(out) can be determined inaccordance with, for example, V(t−t′) in the foregoing expression forcalculating the advance angle δ. That is, the delay width can bedetermined in accordance with a certain previous voltage measurementvalue used among the voltage measurement values of the smoothingcapacitor SC (voltages of the rectified output V_(out)).

As in the first embodiment, the controller 19 may determine the time t′which is a specific deviation width between the variation in the advanceangle δ and the variation in the rectified output V_(out) based on an ACperiod of the AC input V_(in). Thus, even when the AC period of the ACinput V_(in) is changed, it is possible to maintain an optimum value ofthe deviation of the variation of the advance angle δ with respect tothe variation in the rectified output V_(out).

Further, as illustrated in FIG. 23, a timing at which the advance angleδ calculated in this way becomes the second advance angle δ2 is includedin a period in which the voltage of the rectified output V_(out)increases from the minimum value V_(rmin) to the maximum value V_(rmax).

Thus, during the charging period of the smoothing capacitor SC, thestart timing of the power supply period T_(ps) can be advanced from atiming at which the rotation angle of the rotor becomes a predeterminedangle, and more power and current can be supplied from the rectifiercircuit 15 to the three-phase brushless motor M.

On the other hand, in a period other than the period in which thevoltage of the rectified output V_(out) increases from the minimum valueV_(rmin) to the maximum value V_(rmax), the advance angle δ is almostconstant as the first advance angle δ1, that is, the basic advance angleδ_(b). Thus, during a discharging period of the smoothing capacitor SC,it is possible to reduce a discharge amount of the smoothing capacitorSC while maintaining high rotation efficiency of the rotor of thethree-phase brushless motor M.

By reducing inflow and outflow of a current to and from the smoothingcapacitor SC, it is possible to reduce heating of the smoothingcapacitor SC caused due to a capacitor ripple current and improvereliability of the circuit. As the ripple current is less, anelectrolytic capacitor with smaller capacitance can be used. Therefore,it is possible to reduce manufacturing cost of the circuit.

2-3. Experiment Results

Hereinafter, experiment results for verifying results of varying theadvance angle δ with respect to a variation in a voltage of thesmoothing capacitor SC and delaying a variation in the advance angle δby a predetermined time with respect to a variation in the voltage ofthe smoothing capacitor SC will be described.

First, to verify the effect of varying the advance angle δ with respectto the variation in the voltage of the smoothing capacitor SC, resultsobtained by measuring a load current flowing in the U-phase of thethree-phase brushless motor M when the advance angle δ is constant asthe basic advance angle δ_(b) and when the advance angle δ is varied andthe variation is not delayed in the driving device 100 in the drivingdevice 100 are illustrated in FIG. 24.

In the lower drawing of FIG. 24, a plot of a dotted line indicates anabsolute value of a load current when the advance angle δ is not variedand a plot of a solid line indicates an absolute value of a load currentwhen the advance angle δ is varied and the variation is not delayed.

As illustrated in FIG. 24, by varying the advance angle δ with respectto the variation in the voltage of the smoothing capacitor SC, it ispossible to reduce a load current near time t3″ at which the voltage ofthe smoothing capacitor SC particularly becomes the maximum valueV_(rmax) compared to when the advance angle δ is not varied.

Next, the effect of delaying the variation in the advance angle δ by thepredetermined time with respect to the variation in the voltage of thesmoothing capacitor SC will be described. To verify the effect, in thefollowing experiment, the time t′ which is a delay width of a timing atwhich the advance angle δ becomes the second advance angle δ2 withrespect to a timing at which the voltage of the smoothing capacitor SCbecomes the minimum value V_(rmin) was set variously and a load currentwas measured.

First, how the load current changes depending on whether a variation inthe advance angle δ is delayed by the predetermined time with respect toa variation in the voltage of the smoothing capacitor SC will bedescribed with reference to FIG. 25.

In the lower drawing of FIG. 25, a plot of a dotted line indicates anabsolute value of a load current when the variation in the advance angleδ is not delayed by the predetermined time with respect to the variationin the voltage of the smoothing capacitor SC and a plot of a solid lineindicates an absolute value of a load current when the variation in theadvance angle δ is delayed by the predetermined time with respect to thevariation in the voltage of the smoothing capacitor SC.

As illustrated in FIG. 25, by delaying the variation in the advanceangle δ with respect to the variation in the voltage of the smoothingcapacitor SC, a larger load current flows within a period in which thevoltage of the smoothing capacitor SC increases from the minimum valueV_(rmin) to the maximum value V_(rmax) than when the variation in theadvance angle δ is not delayed. That is, by delaying the advance angleδ, the current can flow to the load LO more actively during the chargingperiod of the smoothing capacitor SC.

Next, how a current (ripple current) flowing in and out of the smoothingcapacitor SC is changed when the time t′ at which the variation in theadvance angle δ is delayed with respect to the variation in the voltageof the smoothing capacitor SC is changed variously will be describedwith reference to FIG. 26.

In FIG. 26, the time t′ at which the variation in the advance angle δ isdelayed with respect to the variation in the voltage of the smoothingcapacitor SC was set to 0, t1′″, t2′″, and t3′″(t1′″<t2′″<t3′″). Theripple current was set to an RMS value of the current flowing in and outof the smoothing capacitor SC. Further, the magnitude of the ripplecurrent when the advance angle δ is not varied is indicated by a one-dotchain line as a reference.

As illustrated in FIG. 26, when the advance angle δ is varied and thevariation in the advance angle δ is not delayed with respect to thevariation in the voltage of the smoothing capacitor SC, a ripple currentis slightly larger than when the advance angle δ is not varied. On theother hand, by delaying the variation in the advance angle δ withrespect to the variation in the voltage of the smoothing capacitor SC,the ripple current is less than when the advance angle δ is not varied.The ripple current linearly decreases in a time of delay of thevariation in the advance angle δ with respect to the variation in thevoltage of the smoothing capacitor SC.

That is, by delaying the variation in the advance angle δ with respectto the variation in the voltage of the smoothing capacitor SC, a currentamount flowing in and out of the smoothing capacitor SC is reduced.Thus, it is possible to reduce heating inside the smoothing capacitor SCand improve the reliability. Alternatively, since the capacitance of thesmoothing capacitor SC can be reduced, cost can be reduced.

Further, how a maximum value of the absolute value of the load current,that is, a peak current, is changed when the time t′ at which thevariation in the advance angle δ is delayed with respect to thevariation in the voltage of the smoothing capacitor SC is changedvariously will be described with reference to FIG. 27.

As illustrated in FIG. 27, the maximum value of the absolute value ofthe load current is further reduced by varying the advance angle δ thanwhen the advance angle δ is not varied. When a delay time of thevariation in the advance angle δ with respect to the variation in thevoltage of the smoothing capacitor SC is equal to or greater than thecertain predetermined time t2′″, the maximum value of the load currenttends to increase. Conversely, by setting the delay time of thevariation in the advance angle δ with respect to the variation in thevoltage of the smoothing capacitor SC to be within a predetermined timerange, it is possible to suppress an excessive load current.

From the above-described results, it can be understood that the currentamount flowing in and out of the smoothing capacitor SC is reduced whilesuppressing the excessive current of the three-phase brushless motor Mby varying the advance angle δ and setting the delay time of thevariation in the advance angle δ with respect to the variation in thevoltage of the smoothing capacitor SC to be within the predeterminedrange.

OTHER EMBODIMENTS

As described above, the foregoing embodiments have been described asexamples of technologies disclosed in the present specification.However, the technology in the disclosure is not limited thereto andchanges, substitutions, additions, omissions, and the like can beappropriately made. Accordingly, other embodiments will be exemplifiedbelow.

[1]

The procedures and/or processing content of the processes of theflowcharts illustrated in FIG. 7, FIG. 21, and FIG. 22 can beappropriately changed within the scope of the technology in thedisclosure. For example, a procedure of the measurement of the voltageof the smoothing capacitor SC and the calculation of the duty ratioand/or the advance angle δ in the flowcharts may be reversed. That is,the calculation of the duty ratio and/or the advance angle δ may firstbe performed. Thereafter, the measurement of the voltage of thesmoothing capacitor SC may be performed.

[2]

The foregoing first to third embodiments can be combined. That is, thecontroller 19 may calculate the modulation duty ratio DR at which atleast a period in which the modulation width of the modulation dutyratio DR becomes the maximum value is included in the period from a timeat which the input current input to the smoothing capacitor SC isgenerated to a time at which the voltage of the smoothing capacitor SCbecomes the maximum value V_(rmax) and the advance angle δ at which atleast a period in which the advance angle δ becomes the second advanceangle δ2 which is the maximum value of the variation is included in thatperiod.

In this case, the controller 19 may switch between delay of only themodulation duty ratio DR with respect to the variation in the voltage ofthe smoothing capacitor SC and delay of only the advance angle δ ordelay of both the modulation duty ratio DR and the advance angle δ inaccordance with a predetermined condition.

The controller 19 may change the extent of the effect of delaying themodulation duty ratio DR with respect to the variation in the voltage ofthe smoothing capacitor SC and the extent of the effect of delaying theadvance angle δ in accordance with a predetermined condition. Forexample, by changing the values of the constants A to E of theabove-described expressions for calculating the modulation duty ratio DRand the advance angle δ in accordance with a predetermined condition, itis possible to achieve a change in the extent of the effect.

[3]

In the foregoing first to third embodiments, as described above, themodulation duty ratio DR and the advance angle δ are calculated based onthe previous measurement value of the voltage of the smoothing capacitorSC the predetermined time before. The disclosure is not limited theretoand the modulation duty ratio DR and the advance angle δ can also becalculated based on another parameter.

For example, the controller 19 may periodically vary the modulation dutyratio DR and/or the advance angle δ in a triangular waveform and may setthe timing at which the modulation duty ratio DR and/or the advanceangle δ of the triangular waveform becomes the maximum value to a timinglater than a timing at which the absolute value of the voltage of therectified output V_(out) becomes a minimum.

Specifically, for example, as illustrated in FIG. 28, the controller 19may set the timing of the maximum value of the variation width of themodulation duty ratio DR and/or the advance angle δ varied periodicallyin the triangular waveform to a timing t_(ave) at which the voltage ofthe smoothing capacitor SC becomes an average voltage V_(ave) or atiming slightly delayed from the timing t_(ave).

An increase ratio and/or a decrease ratio of the modulation width of themodulation duty ratio DR and/or the advance angle δ over time in thevariation in the modulation duty ratio DR and/or the advance angle δ ofthe triangular waveform can be adjusted appropriately in accordance witha predetermined condition.

A period in which the modulation width of the modulation duty ratio DRand/or the advance angle δ is the minimum value of the modulation widthof the modulation duty ratio DR and/or the first advance angle δ1 by apredetermined length in the variation in the modulation width of themodulation duty ratio DR and/or the advance angle δ of the triangularwaveform may continue.

[4]

The controller 19 may set the modulation width of the modulation dutyratio DR and/or the advance angle δ to the maximum value of themodulation width of the modulation duty ratio DR and/or the secondadvance angle δ2 from a time at which the absolute value of the voltageof the rectified output V_(out) becomes the first voltage V1 set nearthe minimum value V_(rmin) of the absolute value of the voltage of therectified output to a time at which the absolute value of the voltage ofthe rectified output V_(out) becomes the second voltage V2 set near themaximum value V_(rmax) of the absolute value of the voltage of therectified output V_(out).

Specifically, for example, as illustrated in FIG. 29, the controller 19may set the modulation width of the modulation duty ratio DR and/or theadvance angle δ to the maximum value of the modulation duty ratio DRand/or the second advance angle δ2 during a period of a timing t11 atwhich the voltage of the smoothing capacitor SC becomes the firstvoltage V1 in a period in which the voltage of the smoothing capacitorSC decreases and a timing t12 at which the voltage of the smoothingcapacitor SC becomes the second voltage V2 in a period in which thevoltage of the smoothing capacitor SC increases, and may set themodulation width of the modulation duty ratio DR and/or the advanceangle δ to the minimum value of the modulation duty ratio DR and/or thefirst advance angle δ1 during the other period.

[5]

Further, as illustrated in FIG. 30, the controller 19 may set themodulation duty ratio DR and/or the advance angle δ to the maximum valueof the modulation width of the modulation duty ratio DR and/or thesecond advance angle δ2 during a period of a timing t13 at which thevoltage of the smoothing capacitor SC becomes the first voltage V1 in aperiod in which the voltage of the smoothing capacitor SC increases andthe timing t12 at which the voltage of the smoothing capacitor SCbecomes the second voltage V2 in a period in which the voltage of thesmoothing capacitor SC increases, and may set the modulation duty ratioDR and/or the advance angle δ to the minimum value of the modulationwidth of the modulation duty ratio DR and/or the first advance angle δ1during the other period.

In this case, as illustrated in FIG. 30, the modulation duty ratio DRand/or the advance angle δ becomes the maximum value of the modulationwidth of the modulation duty ratio DR and/or the second advance angle δ2during the period in which the absolute value of the voltage of therectified output V_(out) increases.

The modulation duty ratio DR and/or the advance angle δ that becomes themaximum value of the modulation width of the modulation duty ratio DRand/or the second advance angle δ2 during the period in which theabsolute value of the voltage of the rectified output V_(out) increasescan also be calculated by setting the modulation width of the modulationduty ratio DR and/or the advance angle δ to the maximum value of themodulation duty ratio DR and/or the second advance angle δ2, forexample, when a current measurement value of the voltage of thesmoothing capacitor SC is greater than a previous measurement value nearthe current measurement value.

According to the methods described in [3] to [5], the controller 19 cancalculate the modulation duty ratio DR and/or the advance angle δ atwhich at least a portion of the period in which the variation width ofthe modulation duty ratio DR and/or the advance angle δ becomes themaximum value of the variation width of the modulation duty ratio DRand/or the second advance angle δ2 is included in the period from thetime at which the input current input to the smoothing capacitor SC isgenerated to the time at which the voltage of the smoothing capacitor SCbecomes the maximum value V_(rmax).

[6]

When the switching circuit 17 is particularly a converter such as acritical-mode boosting chopper type converter, a critical-mode step-downchopper type converter, an LLC converter, or a pseudo-resonant flybackconverter, the controller 19 may vary the switching period T_(sw)between a first period T1 _(sw) and a second period T2 _(sw) greaterthan the first period T1 _(sw).

Specifically, the controller 19 may set the switching period T_(sw) suchthat at least a portion of a period in which the switching period T_(sw)becomes a second period is included in a period from the time at whichthe input current input to the smoothing capacitor SC is generated tothe time at which the voltage of the smoothing capacitor SC becomes themaximum value V_(rmax). The variation in the switching period T_(sw) canbe realized similarly to the above-described method of varying themodulation duty ratio DR and the advance angle δ.

Thereafter, the controller 19 can performs a switching operation of theswitching circuit 17 by switching between the ON-state and the OFF-stateof the switching circuit 17 at the switching period T_(sw) set in thisway and varying the switching period T_(sw) between the first period T1_(sw) and the second period T2 _(sw) greater than the first period T1_(sw).

[7]

The controller 19 may detect a period from a time at which the inputcurrent input to the smoothing capacitor SC is generated to a time atwhich the voltage of the smoothing capacitor SC become the maximum valueV_(rmax) based on the voltage value and/or the phase of the AC inputV_(in).

The input current with which the smoothing capacitor SC is charged isgenerated when the absolute value of the voltage of the AC input V_(in)is greater than the voltage of the smoothing capacitor SC. The voltageof the smoothing capacitor SC increases slightly later due to aninfluence of an inductor component or the like in the circuit.

After the absolute value of the voltage of the AC input V_(in) becomesthe maximum value (the phase of the AC input V_(in) is (2π+1)π/2 (wheren is an integer)), the decrease in the input current starts near attiming at which the absolute value of the voltage of the AC input V_(in)decreases and becomes less than the voltage of the smoothing capacitorSC, and the input current finally becomes 0. The voltage of thesmoothing capacitor SC starts to decrease slightly before the decreasein the input current starts.

Accordingly, the controller 19 can detect a period from a time at whichthe input current input to the smoothing capacitor SC is generated to atime at which the voltage of the smoothing capacitor SC becomes themaximum value V_(rmax) based on the voltage value and/or the phase ofthe AC input V_(in) in consideration of a factor causing deviationbetween a timing at which the input current is generated and decreasesand a timing at which the voltage of the smoothing capacitor SCincreases and decreases.

Examples of the factor causing the deviation between the timing at whichthe input current is generated and decreases and the timing at which thevoltage of the smoothing capacitor SC increases and decreases include aninductor component in the circuit, power to the load LO, and a ripplevoltage of the smoothing capacitor SC.

[8]

When the start timing of the power supply period T_(ps) is adjusted inthe foregoing third embodiment, an end timing of the power supply periodT_(ps) is also moved with adjustment of the start timing. That is, evenwhen the start timing is adjusted, the length of the power supply periodT_(ps) is constant.

However, the present invention is not limited thereto and the controller19 may adjust the start timing by increasing or decreasing the powersupply period T_(ps). Adjustment of the start timing by increasing ordecreasing the power supply period T_(ps) is also referred to asconduction angle modulation.

For example, by fixing the end timing of the power supply period T_(ps)before and after adjustment of the start timing and increasing the powersupply period T_(ps), the start timing can be moved temporally backward,that is, the advance angle δ can be increased. On the other hand, byfixing the end timing of the power supply period T_(ps) before and afteradjustment of the start timing and decreasing the power supply periodT_(ps), the start timing can be moved temporally forward, that is, theadvance angle δ can be decreased.

In addition, for example, by fixing a predetermined timing in the powersupply period T_(ps) before and after adjustment of the start timing andincreasing the power supply period T_(ps), it is possible to move thestart timing temporally backward and move the end timing temporallyforward. On the other hand, by fixing the predetermined timing in thepower supply period T_(ps) before and after adjustment of the starttiming and decreasing the power supply period T_(ps), it is possible tomove the start timing temporally forward and move the end timingtemporally backward.

[9]

In the foregoing first to third embodiments, the PWM control of theswitching elements SW1 to SW3 of the switching circuit 17 which is aninverter is performed. On the other hand, the switching elements SW4 toSW6 repeat a low-speed switching operation in which the ON-statecontinues for only T/3 and the OFF-state continues for only 2T/3, asillustrated in FIG. 6 and FIG. 20.

However, the present invention is not limited thereto. While performingthe PWM control of the switching elements SW4 to SW6, the switchingelements SW1 to SW3 may be caused to perform the switching operation ata low speed as in the switching elements SW4 to SW6 in the first tothird embodiments.

[10]

In the foregoing first to third embodiments, for power factorimprovement of the input current, a power factor improvement circuitincluding the inductor element L and the smoothing capacitor SC, aso-called passive power factor correction (PFC) circuit, is used.However, a power factor improvement circuit including a switchingcircuit such as a chopper circuit, a so-called active PFC circuit, maybe used. In this case, a higher power factor can be obtained and aripple current generated with an AC period can be reduced, and thus itis possible to obtain the advantage of reducing the cost because ofsuppression of deterioration and miniaturization of the electrolyticcapacitor.

REFERENCE SIGNS LIST

-   -   100 Driving device    -   1 Power supply device    -   11 Input unit    -   I1 First input terminal    -   I2 Second input terminal    -   13 Output unit    -   15 Rectifier circuit    -   151 Rectifier unit    -   L Inductor element    -   SC Smoothing capacitor    -   17 Switching circuit    -   SW1 to SW6 Switching element    -   D1 to D10 Rectifier element    -   19 Controller    -   191 Measurement unit    -   LO Load    -   M Three-phase brushless motor    -   PS AC power source    -   DR Duty ratio    -   DR_(B) Basic duty ratio    -   DR1 First duty ratio    -   DR2 Second duty ratio    -   T_(sw) Switching period    -   T1 _(sw) First period    -   T2 _(sw) Second period    -   T_(ps) Power supply period    -   V1 First voltage    -   V2 Second voltage    -   V_(in) AC input    -   V_(out) Rectified output    -   V_(ave) Average voltage    -   V_(rmax) Maximum value    -   V_(rmin) Minimum value    -   δ Advance angle    -   δ1 First advance angle    -   δ2 Second advance angle

1. A power supply device comprising: an output unit configured toconnect a load; an input unit configured to input an AC input varying ata predetermined period between a positive voltage and a negativevoltage; a rectifier circuit configured to convert the AC input from theinput unit into a rectified output which is one of the positive voltageand a negative voltage and include a smoothing capacitor that smoothesthe rectified output; a switching circuit configured to connect thesmoothing capacitor as an input, connect the output unit as an output,and switch between an ON-state in which an input impedance viewed fromthe smoothing capacitor is low and an OFF-state in which an inputimpedance is higher than the input impedance in the ON-state at aswitching period shorter than the predetermined period; and a controllerconfigured to set a modulation duty ratio such that at least a portionof a period in which a modulation width of the modulation duty ratiobecomes a maximum value is included in a period from a time at which aninput current input to the smoothing capacitor is generated to a time atwhich a voltage of the smoothing capacitor becomes a maximum when theswitching circuit is controlled to switch between the ON-state and theOFF-state while outputting the modulation duty ratio at which a dutyratio which is a ratio of a period in which the ON-state is maintainedto the switching period is modulated in accordance with a variation inthe rectified output.
 2. The power supply device according to claim 1,wherein the controller includes a measurement unit that measures avoltage of the rectified output.
 3. The power supply device according toclaim 1, wherein the smoothing capacitor is an electrolytic capacitor.4. The power supply device according to claim 1, wherein the controllersets the modulation duty ratio such that at least the portion of theperiod in which the modulation width of the modulation duty ratiobecomes the maximum value is included in a period in which an absolutevalue of a voltage of the rectified output increases.
 5. The powersupply device according to claim 1, wherein the controller sets amodulation width of the modulation duty ratio as a periodic variationdeviating by a predetermined time from a periodic variation of therectified output.
 6. The power supply device according to claim 5,wherein the predetermined time is determined based on a frequency of theAC input.
 7. The power supply device according to claim 5, wherein thecontroller sets the modulation width of the modulation duty ratio basedon a reciprocal of an absolute value of a previous voltage of therectified output before the predetermined time.
 8. The power supplydevice according to claim 5, wherein the controller periodically variesthe modulation width of the modulation duty ratio in a triangularwaveform and sets a timing at which the modulation width of themodulation duty ratio of a triangular waveform becomes a maximum valueto a timing later than a timing at which an absolute value of a voltageof the rectified output becomes a minimum.
 9. The power supply deviceaccording to claim 1, wherein the controller sets the modulation widthof the modulation duty ratio to the maximum value from a time at whichan absolute value of a voltage of the rectified output becomes a firstvoltage set near a minimum value of the absolute value of the voltage ofthe rectified output to a time at which an absolute value of a voltageof the rectified output becomes a second voltage set near a maximumvalue of the absolute value of the voltage of the rectified output. 10.The power supply device according to claim 1, wherein the controllersets the modulation width of the modulation duty ratio to the maximumvalue during a period in which an absolute value of a voltage of therectified output increases.
 11. The power supply device according toclaim 1, wherein the switching circuit is an inverter circuit.
 12. Thepower supply device according to claim 1, wherein the controller setsthe modulation duty ratio based on a difference between a target poweroutput to the output unit and an actual power output to the output unit.13. The power supply device according to claim 1, wherein the controllercalculates the modulation duty ratio based on a basic duty ratio setindependently from a variation in the rectified output and the variationof the rectified output.
 14. The power supply device according to claim13, wherein the basic duty ratio is set based on a difference between atarget power output to the output unit and an actual power output to theoutput unit.
 15. The power supply device according to claim 13, whereinthe basic duty ratio varies at a predetermined period.
 16. The powersupply device according to claim 13, wherein the switching circuitincludes a plurality of switching elements having mutually differentphases and switches between the ON-state and the OFF-state at theswitching period, and wherein the basic duty ratio is set as a ratio ofa period in which the ON-state of each switching element is maintainedto the switching period independently from the variation in therectified output.
 17. The power supply device according to claim 16,wherein the basic duty ratio of each switching element is set based on adifference between a target power output to the output unit and anactual power output to the output unit.
 18. The power supply deviceaccording to claim 16, wherein the basic duty ratio of each switchingelement varies at a predetermined period of which a phase is differentfrom the basic duty ratio of the other switching elements.
 19. The powersupply device according to claim 18, wherein the controller modulatesthe basic duty ratio of each switching element by a two-phase modulationscheme.
 20. A driving device comprising: the power supply deviceaccording to claim 1; and a motor connected to the output unit.
 21. Thedriving device according to claim 20, wherein the controller sets themodulation duty ratio based on a difference between a target rotationspeed of the motor and an actual rotation speed of the motor.
 22. Apower supply device comprising: an output unit configure to connect aload; an input unit configured to input an AC input varying at apredetermined period between a positive voltage and a negative voltage;a rectifier circuit configured to convert the AC input from the inputunit into a rectified output which is one of the positive voltage andthe negative voltage and include a smoothing capacitor that smoothes therectified output; a switching circuit configured to connect thesmoothing capacitor as an input, connect the output unit as an output,and switch between an ON-state in which an input impedance viewed fromthe smoothing capacitor is low and an OFF-state in which an inputimpedance is higher than the input impedance in the ON-state at aswitching period shorter than the predetermined period; and a controllerconfigured to set the switching period such that at least a portion of aperiod in which the switching period is a second period is included in aperiod from a time at which an input current input to the smoothingcapacitor is generated to a time at which a voltage of the smoothingcapacitor becomes a maximum when the switching circuit is controlled toswitch between the ON-state and the OFF-state by varying the switchingperiod between a first period and the second period greater than thefirst period.
 23. A control method for a power supply device including:an output unit that connects a load, an input unit that inputs an ACinput varying at a predetermined period between a positive voltage and anegative voltage, and a rectifier circuit that converts the AC inputfrom the input unit into a rectified output which is one of the positivevoltage and the negative voltage and includes a smoothing capacitor thatsmoothes the rectified output, and a switching circuit that connects thesmoothing capacitor as an input, connects the output unit as an output,and switches between an ON-state in which an input impedance viewed fromthe smoothing capacitor is low and an OFF-state in which an inputimpedance is higher than the input impedance in the ON-state at aswitching period shorter than the predetermined period, the controlmethod comprising steps of: setting a modulation duty ratio such that atleast a portion of a period in which a modulation width of themodulation duty ratio becomes a maximum value is included in a periodfrom a time at which an input current input to the smoothing capacitoris generated to a time at which a voltage of the smoothing capacitorbecomes a maximum when outputting a modulation duty ratio at which aduty ratio which is a ratio of a period in which the ON-state ismaintained to the switching period is modulated in accordance with avariation in the rectified output; and controlling the switching betweenthe ON-state and the OFF-state of the switching circuit by adjusting alength in which the ON-state is maintained and a length in which theOFF-state is maintained in accordance with the set modulation dutyratio.
 24. The control method according to claim 23, wherein the step ofsetting the modulation duty ratio includes a step of calculating themodulation duty ratio based on a basic duty ratio set independently fromthe variation in the rectified output and the variation of the rectifiedoutput.
 25. The control method according to claim 24, wherein theswitching circuit includes a plurality of switching elements havingmutually different phases and switches between the ON-state and theOFF-state at the switching period, and wherein the basic duty ratio isset as a ratio of a period in which the ON-state of each switchingelement is maintained to the switching period independently from thevariation in the rectified output.
 26. The control method according toclaim 25, wherein, in the step of setting the modulation duty ratio, thebasic duty ratio of each switching element is set based on a differencebetween a target power output to the output unit and an actual poweroutput to the output unit.
 27. The control method according to claim 25,wherein the basic duty ratio of each switching element varies at apredetermined period of which a phase is different from the basic dutyratio of the other switching elements.
 28. The control method accordingto claim 27, wherein, in the step of setting the modulation duty ratio,the basic duty ratio is modulated by a two-phase modulation scheme. 29.A control method for a power supply device including an output unit thatconnects a load, an input unit that inputs an AC input varying at apredetermined period between a positive voltage and a negative voltage,a rectifier circuit that converts the AC input from the input unit intoa rectified output which is one of the positive voltage and the negativevoltage and includes a smoothing capacitor that smoothes the rectifiedoutput, and a switching circuit that connects the smoothing capacitor asan input, connects the output unit as an output, and switches between anON-state in which an input impedance viewed from the smoothing capacitoris low and an OFF-state in which an input impedance is higher than theinput impedance in the ON-state at a switching period shorter than thepredetermined period, the control method comprising steps of: settingthe switching period such that at least a portion of a period in whichthe switching period is a second period is included in a period from atime at which an input current input to the smoothing capacitor isgenerated to a time at which a voltage of the smoothing capacitorbecomes a maximum when the switching circuit is controlled to switchbetween the ON-state and the OFF-state by varying the switching periodbetween a first period and the second period greater than the firstperiod; and switching between the ON-state and the OFF-state of theswitching circuit at the set switching period.
 30. A storage mediumstoring a program, wherein the program causes a computer to perform thecontrol method according to claim 23.